Semiconductor memory device and defect remedying method thereof

ABSTRACT

A semiconductor memory device formed on a semiconductor chip includes first memory arrays, a plurality of second memory arrays, a first voltage generator, and first bonding pads. The semiconductor chip is divided into first, second and third rectangle regions and the third rectangle region is arranged between the first rectangle region and the second rectangle region. The first memory arrays are formed in the first rectangle region. The second memory arrays are formed in the second rectangle region. The voltage generator and first bonding pads are arranged in the third rectangle region. The first bonding pads are arranged between the first rectangle region and the voltage generator and no bonding pads are arranged between the voltage generator and the second memory arrays.

CROSS-REFERENCE TO RELATED APPLICATIONS

This Application is a continuation of U.S. application Ser. No.11/330,220, filed Jan. 12, 2006, which, in turn, is a continuation ofU.S. application Ser. No. 11/101,504, filed Apr. 8, 2005 (now U.S. Pat.No. 7,016,236), which, in turn, is a continuation of U.S. applicationSer. No. 10/683,260 (now U.S. Pat. No. 6,898,130), filed Oct. 14, 2003;which, in turn, is a continuation of application Ser. No. 10/254,980,filed on Sep. 26, 2002 (now U.S. Pat. No. 6,657,901), which is acontinuation of application Ser. No. 10/000,032, filed on Dec. 4, 2001(now U.S. Pat. No. 6,515,913), which is a continuation of applicationSer. No. 09/714,268, filed on Nov. 17, 2000 (now U.S. Pat. No.6,335,884) which is a continuation of application Ser. No. 09/547,917,filed on Apr. 11, 2000 (now U.S. Pat. No. 6,212,089), which is acontinuation of application Ser. No. 09/361,203, filed on Jul. 27, 1999(now U.S. Pat. No. 6,160,744), which is a continuation of applicationSer. No. 08/618,381, filed on Mar. 19, 1996 (now U.S. Pat. No.5,854,508), which is a continuation of Ser. No. 08/455,411, filed on May31, 1995 (now U.S. Pat. No. 5,579,256) which is a continuation of Ser.No. 08/159,621, filed on Dec. 1, 1993 (now U.S. Pat. No. 5,602,771)which is a division of Ser. No. 07/899,572, filed on Jun. 18, 1992 nowabandoned which is a continuation of Ser. No. 07/424,904, filed on Oct.18, 1989 (now abandoned), the entire disclosures of which are herebyincorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a semiconductor device and its defectremedying method and, more particularly, to a technology which iseffective if applied to a dynamic RAM (Random Access Memory) having astorage capacity as large as about 16 Mbits.

2. Description of the Prior Art

The development of the dynamic RAM having the large storage capacity ofabout 16 Mbits is being advanced. An example of the dynamic RAM isdescribed on pp. 67 to 81 of “Nikkei Microdevice” issued on Mar. 1, 1988by NIKKEI McGRAW-HILL.

In accordance with the increase in the storage capacity, the memory chipnecessarily has its size enlarged. Accordingly, special considerationshave to be taken into the drop of the operation speed, which is causedby making the elements finer and by handling the wiring lines. In otherwords, the realization of the high storage capacity of about 16 Mbitsrequires development of a new technology which is different from thatused for the dynamic RAM of about 1 or 4 Mbits.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a semiconductor devicewhich aims at having a large storage capacity.

Another object of the present invention is to provide a semiconductorstorage device which realizes the large storage capacity while speedingup the operations.

Still another object of the present invention is to provide a rationaldefect remedying method for the memory device aiming at the largestorage capacity.

The aforementioned and other objects and novel features of the presentinvention will become apparent from the descriptions to be made in thefollowing with reference to the accompanying drawings.

The representatives of the invention to be disclosed hereinafter will bebriefly described in the following.

There is provided a semiconductor memory device having a large storagecapacity, in which a semiconductor chip is bisected by its longitudinalcenter line to form two regions, in which peripheral circuits arearranged in a cross area composed of the longitudinal center portionsand the transverse center portions of said two regions, and in whichmemory arrays are arranged in the four regions which are divided by saidcross area. In the cross area, the edges contacting with the memoryarrays are arranged with X-decoders and Y-decoders, and the regions ofthe longitudinal or transverse center portion, which are interposedbetween the X-decoders, are arranged with a main amplifier, a commonsource switch circuit, a sense amplifier control signal generator and amat selection control circuit. Those circuits of the peripheralcircuits, which may probably inject minority carriers into a substrateon principle, are arranged on two center lines of the cross area ortheir vicinities. The memory arrays formed in the four quartered areasof said cross area are constructed of a block of plural memory mats as aunit having the same size as includes the sense amplifiers. The unitmemory mat includes a control circuit for generating a variety of timingsignals for the memory cell selections on the basis of a mat selectionsignal. The control circuit is activated by the mat selection signal.The memory mat selection signal is prepared by decoding the addresssignal inputted through a specific address buffer. Bonding pads arepartially or wholly arranged in the regions of said cross area. Thebonding pads are bonded to LOC lead frame. Of these bonding pads, aplurality of pads for applying the power voltage of the circuit and theground potential are arranged at a suitable spacing according to circuitblocks requiring them and are connected to the common LOC lead frame tobe fed with the power voltage of the circuit and the ground potential.The four regions quartered by the cross area are arranged with thememory arrays, and the semiconductor chip has its four corners stepped.There is provided an internal drop voltage generator which is madeoperative in response to the power voltage fed from an external terminaland which includes one or more impedance converting output buffers madereceptive of a reference voltage prepared by a reference voltagegenerator. The internal drop voltage generator is provided for each ofthe memory array operating voltage and the peripheral circuit operatingvoltage. The internal drop voltage generator drives an output MOSFET ofsource-follower type having its gate fed with the signal to be outputtedthrough a level converter for converting the signal to be fed and formedby the internal circuit into a signal level corresponding to the powervoltage fed from the external terminal. The drop voltage generated bythe internal drop voltage generator is selectively outputted, in theoutput high impedance state of a data output buffer in a test mode, fromthe output terminal of the data output buffer through a switch MOSFET tobe switched by a signal at the bootstrap voltage or external powervoltage level. The selection signal of the word lines or the sharedsense amplifiers is prepared by a selector which has its operationscontrolled by a high voltage prepared by boosting the internal dropvoltage. At least one pair of memory cell arrays are arrangedsymmetrically with respect to the main amplifier, and the main amplifieris selectively connected with the input/output lines of the pairedmemory cell arrays through a switch circuit to be switched in accordancewith the selections of the paired memory cell arrays. The shared senseamplifier is given an operation mode for connecting both the data linesat the selected and unselected sides. The pull-up MOSFET of CMOSstructure composed of the sense amplifier, the initial-step circuit ofthe main amplifier and the input/output lines, the short MOSFET composedof the complementary data lines and the complementary input/outputlines, and the MOSFET of diode mode constituting the charge pump circuitare caused to have a low threshold voltage. A pair of parallel bit linesare constructed of the bit line cross type, in which the bit lines areinterchanged by using a first metal wiring layer formed over the wiringlayer forming the bit lines. The first metal wiring layer also forms thecolumn selection lines, one of which is formed to correspond two pairsof bit lines and folded to overlap from one to other bit line pair atportion different from the cross portion of the bit lines. A stepdamping region made of a dummy wiring layer is formed between a memorycell array of laminated type and a peripheral circuit.

There is also provided a defect remedying method comprising the stepsof: constructing a memory array of a block composed of plurality unitsof memory mats having the same size and including sense amplifiers;forming redundancy word lines and/or redundancy data lines for each ofsaid memory mats; forming redundancy decoders of a number smaller thanthe total number of the redundancy word and/or data lines of all of saidmemory mats and larger than the total number of the redundancy wordand/or data lines of each of said memory mats so that said redundancydecoders may be used for each of said memory mats or commonly for saidmemory mats. Preparatory word lines and/or preparatory column selectionlines wired to intersect a plurality of word lines and/or columnselection lines, respectively, are formed at the output of a word lineor column selector, and the word lines and/or the output lines of thecolumn selector are cut by physical means, when a word line and/or adata line are defective, from the column selection lines correspondingto the defective word line and/or the defective data line and areconnected with the preparatory word lines and/or the preparatory columnselection lines. When in the multi-bit simultaneous testing mode by themultiplex selection of the column system, only the defective data orcolumn selection line of the data lines or column selection lines isswitched to a redundancy data line or a redundancy column selection in amanner to correspond to the memory cell array divided into a pluralityof memory blocks. The data lines are divided into a plurality of blocksby one of a specific-bit of the address signals of the row and/or columnsystems, a block address prepared inside, or the combination of theaddress signal and the block address, and a defective data line in adefective block only is switched to a redundancy data line by making useof a signal designating the block.

Since the major timing signals are propagated four ways from the centerof the chip, according to the means specified above, the lengths of thesignal wiring lines, which might otherwise accompany the size-up of thechip, can be substantially shortened to realize the large capacity andthe speed-up of the DRAM. The influences to be exerted upon the memoryarrays can be minimized by arranging a circuit capable of generatingminority carriers on or in the vicinity of the two center lines of theaforementioned cross area. The design and control can be simplified bymaking a block of the unit memory mats of the same size containing senseamplifiers. The bonding pads are connected with the LOC lead frame sothat their arrangement can be optimized. Thanks to the provision of thepads for establishing the circuit power voltage and the groundpotential, the power impedance can be dropped. The stress from the resinmold can be dispersed by the steps formed in the corners. Thanks to theprovision of the internal drop voltage generator, it is possible toprevent the electrostatic breakdown which might otherwise be caused as aresult of the low power consumption and the finer element. Since thevoltage is dropped in a manner to correspond to the memory arrayoperating voltage and the peripheral circuit operating voltage, it ispossible to increase the power noise margin. The output level isretained and speeded up by driving the output MOSFET with the levelchange. The internal voltage can be monitored by bringing the dataoutput buffer to the output high-impedance state. The speed-up and thestabilization can be achieved by causing the boost power to form theselection signals for selecting the word lines and the shared senseamplifiers. The circuit can be simplified by causing the main amplifiersto correspond to the plural memory cell arrays. The margin test of thesense amplifiers can be executed by connecting the shared senseamplifiers with the two data lines. The speed-up and the level drop canbe minimized by using the MOSFET having a low threshold voltage. Thehigh integration can be attained by interchanging the bit lines usingthe metal wiring layer formed over the bit lines. The metal wiring layercan also be used as the column selection line. The stepped shockabsorbing region can prevent the shortage of the steps in the wiringlines.

As the defect remedying method, the redundancy circuit can be simplifiedby utilizing the redundancy decoder as the multiple memory mats. Thecircuit can be simplified and speeded up by switching the defective dataline or word line directly to the preparatory data line or word line.The preparatory circuit can be simplified by interchanging only thedefective circuit when in the multi-bit simultaneous test most by themultiplex selection of the Y-system. Thus, the defect remedy can beachieved with the simple structure by utilizing the block designatingsignals.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a fundamental layout showing one embodiment of the dynamicRAM, to which is applied the present invention;

FIG. 2 is an overall layout showing one embodiment of the DRAM accordingto the present invention;

FIG. 3 is a layout showing the detailed arrangement of bonding pads ofthe DRAM;

FIG. 4 is a block diagram showing one embodiment of the addressassignment of the same;

FIG. 5 is a block diagram noting the control signals of the dynamic RAMaccording to the present invention;

FIG. 6 is a block diagram showing the operation sequence of the dynamicRAM according to the present invention;

FIG. 7 is a layout for specifically explaining the power supply linesand the relations between the associated power circuits and the pads;

FIG. 8 is a layout for specifically explaining the ground lines and therelations between the associated internal power circuits and the pads;

FIGS. 9(A) and 9(B) are a specific layout and a section showing oneembodiment of an input protection circuit according to the presentinvention;

FIG. 10 is a specific layout showing one embodiment of the inputprotection circuit mounted in the external power source voltage pad;

FIG. 11 is a layout showing one embodiment of the peripheral portion ofa semiconductor chip;

FIG. 12 is a schematic section showing the corner of the peripheralportion;

FIG. 13 is a schematic section showing the outermost periphery;

FIG. 14 is a fundamental layout showing another embodiment of thedynamic RAM according to the present invention;

FIG. 15 is a fundamental layout showing still another embodiment of thedynamic RAM;

FIG. 16 is a fundamental layout showing a further embodiment of thedynamic RAM;

FIGS. 17(A) to 17(C) are layouts showing the fundamental structure ofanother embodiment of the memory mat and another embodiment of thememory block constructed by combining the former;

FIGS. 18(A) to 18(C) are layouts showing the fundamental structure ofanother embodiment of the aforementioned memory mat and anotherembodiment of the memory block constructed by combining the former;

FIGS. 19(A) to 19(C) are layouts showing the fundamental structure ofanother embodiment of the aforementioned memory mat and anotherembodiment of the memory block constructed by combining the former;

FIGS. 20(A) to 20(C) are layouts showing the fundamental structure ofstill another embodiment of the aforementioned memory mat and anotherembodiment of the memory block constructed by combining the former;

FIGS. 21(a) and 21(B) are layouts showing the fundamental structure ofanother embodiment of the aforementioned memory mat and anotherembodiment of the memory block constructed by combining the former;

FIG. 22 is a top plan view showing one embodiment of the lead frame tobe used in the dynamic RAM according to the present invention;

FIGS. 23(A) to 23(C) are schematic side elevations showing examples ofconnection between the lead frame and the semiconductor chip;

FIGS. 24(A) and 24(B) are external views and an internal perspectiveview showing one embodiment of the dynamic RAM according to the presentinvention;

FIGS. 25(A) to 25(C) are views showing the pin arrangements of externalterminals according to one embodiment of the dynamic RAM of the presentinvention;

FIG. 26 is a layout showing the pin arrangement of the externalterminals according to one embodiment in case a ZIP package is used;

FIG. 27 is a layout showing the pin arrangement of the externalterminals according to one embodiment in case an SOJ package is used;

FIG. 28 presents diagrams showing portions of the circuit according toone embodiment of the RAS control circuit of the dynamic RAM of thepresent invention;

FIG. 29 presents diagrams showing portions of the circuit according toone embodiment of the aforementioned control circuit;

FIG. 30 presents diagrams showing other portions of the circuitaccording to one embodiment of the aforementioned control circuit;

FIG. 31 is a circuit diagram showing one embodiment of the X-addressbuffer of the dynamic RAM according to the present invention;

FIG. 32 is a circuit diagram showing one embodiment of the addressbuffer corresponding to the X-address signals A9 and A10;

FIG. 33 is a circuit diagram showing one embodiment of the addressbuffer corresponding to the X-address signal A11;

FIG. 34 is a circuit diagram showing one embodiment of the addressbuffer corresponding to the X-address signal A8;

FIG. 35 presents circuit diagrams showing portions of one embodiment ofa predecoder of a row system;

FIG. 36 is a circuit diagram showing one embodiment of a redundancycircuit of an X-system;

FIG. 37 presents circuit diagrams showing portions of one embodiment ofa decoder circuit for selecting the word lines;

FIG. 38 presents circuit diagrams showing portions of one embodiment ofa decoder circuit for selecting the redundant word lines;

FIG. 39 is a circuit diagram showing one embodiment of a timinggenerator for activating the sense amplifier;

FIG. 40 presents circuit diagrams showing portions of one embodiment ofa control circuit disposed in the memory mat;

FIG. 1 is a circuit diagram showing one embodiment of an X-decoder, aword line driver and a shared control line driver;

FIG. 42 is a circuit diagram showing one embodiment of a memory cellarray;

FIG. 43 is a circuit diagram showing one embodiment of a refresh addresscounter;

FIG. 44 presents circuit diagrams showing portions of one embodiment ofa control circuit of the CAS system;

FIG. 45 is a circuit diagram showing one embodiment of a Y-addressbuffer;

FIG. 46 presents circuit diagrams showing portions of one embodiment ofa Y-redundancy circuit;

FIG. 47 presents circuit diagrams showing other portions of oneembodiment of a Y-redundancy circuit;

FIG. 48 presents circuit diagrams showing portions of one embodiment ofa Y-redundancy circuit;

FIG. 49 is a circuit diagram showing one embodiment of the predecoder ofa Y-address signal;

FIG. 50 is a circuit diagram showing one embodiment of the Y-decoder forgenerating a column selection signal;

FIG. 51 is a circuit diagram showing one embodiment a nibble counter;

FIG. 52 presents circuit diagrams showing portions of one embodiment ofa control circuit for generating a Y-control signal;

FIG. 53 is a circuit diagram showing one embodiment of the operationmode decoder;

FIG. 54 presents circuit diagrams showing portions of one embodiment ofa Y-control circuit;

FIG. 55 presents circuit diagrams showing portions of one embodiment ofa WE-control circuit;

FIG. 56 presents circuit diagrams showing other portions of oneembodiment of a WE-control circuit;

FIG. 57 is a circuit diagram showing one embodiment of the data inputbuffer;

FIG. 58 is a circuit diagram showing one embodiment of the mainamplifier control circuit;

FIG. 59 is a circuit diagram showing one embodiment of the mainamplifier;

FIG. 60 is a circuit diagram showing one embodiment of the outputcontrol circuit of the data of the main amplifier;

FIG. 61 is a circuit diagram showing one embodiment of the outputcontrol circuit of the main amplifier;

FIG. 62 is a circuit diagram showing one embodiment of the data outputbuffer;

FIG. 63 presents circuit diagrams showing portions of one embodiment ofthe test circuit;

FIG. 64 presents circuit diagrams showing other portions of oneembodiment of the test circuit;

FIG. 65 is a circuit diagram showing one embodiment of the controlcircuit for designating the operation mode;

FIG. 66 is a circuit diagram showing one embodiment of another controlcircuit;

FIG. 67 is a circuit diagram showing one embodiment of the substrateback bias voltage generator;

FIG. 68 is a circuit diagram showing one embodiment of the internalboost voltage generator;

FIG. 69 is a circuit diagram showing one embodiment of the internal dropvoltage generator;

FIG. 70 is a timing chart showing one example of the operations of theRAS system;

FIG. 71 is a timing chart showing one example of the operations of theRAS system;

FIG. 72 is a timing chart showing one example of the operations of theRAS system;

FIG. 73 is a timing chart showing one example of the operations of theX-address buffer;

FIG. 74 is a timing chart showing one example of the operations of theCAS system;

FIG. 75 is a timing chart showing one example of the address selectingoperations of the CAS system;

FIG. 76 is a timing chart showing one example of the writing operations;

FIG. 77 is a timing chart showing one example of the operations of theY-address buffer;

FIG. 78 is a timing chart showing one example of the operations of thetest mode;

FIG. 79 is a timing chart showing one example of the operations of theCAS system;

FIG. 80 is a timing chart showing one example of the operations of theCAS system;

FIG. 81 is a timing chart showing one example of the operations of theCAS system;

FIG. 82 is a block diagram showing another embodiment of the defectremedying method according to the present invention;

FIG. 83 is a block diagram showing another embodiment of the defectremedying method according to the present invention;

FIGS. 84(A) to 84(C) are waveform charts and a circuit diagram of oneembodiment for explaining a word line testing method;

FIGS. 85(A) to 85(D) are waveform charts and circuit diagrams of oneembodiment for explaining a signal amount margin testing method;

FIG. 86 is a block diagram showing another embodiment of the functionset mode;

FIGS. 87(A) to 87(C) are waveform charts and a circuit diagram of oneembodiment showing another embodiment of the refresh address counter;

FIGS. 88(A) and 88(B) a block diagram showing another embodiment of theinternal power monitor method and a waveform chart for explaining thesame;

FIGS. 89(A) and 89(B) are a circuit diagram and a waveform chart forexplaining the principle of the multi-bit testing method;

FIG. 90 is a section showing an element structure taken in the bit linedirection according to one embodiment of the present invention;

FIGS. 91(A) to 91(C) are conceptional diagrams for explaining the defectremedying method according to the present invention;

FIG. 92 is a block diagram showing one embodiment of the layout of themain amplifier and the memory cell array according to the presentinvention;

FIG. 93 is a block diagram showing another embodiment of the layout ofthe main amplifier and the memory cell array according to the presentinvention;

FIG. 94 is a fundamental layout showing another embodiment of thesemiconductor chip according to the present invention;

FIG. 95 is a pattern diagram showing one embodiment of the memory cellarray according to the present invention;

FIGS. 96(A) and 96(B) are a section and a schematic diagram forexplaining the bit line cross portion of the memory cell array;

FIGS. 97 to 99 are pattern diagrams showing one embodiment of the sharedsense amplifier column portion in the bit line direction and thecorresponding memory cell array portion;

FIG. 100 is a section showing the step damping region;

FIG. 101 is a pattern diagram showing one embodiment of the memory cellarray in the word line direction and the corresponding word driver;

FIGS. 102 to 105 are pattern diagrams each showing one embodiment of thecorresponding word drivers;

FIGS. 106 and 107 are pattern diagrams each showing one embodiment ofthe corresponding X-decoder;

FIG. 108 is a pattern diagram showing one embodiment of the memory cellarray in the word line direction and the word clear circuit;

FIG. 109 is a diagram showing one embodiment of the present invention,in which the number of resresh cycles is √{square root over (n)} whereasthe number of sense amplifiers to be simultaneously activated is Vn fora capacity of n bits;

FIG. 110 is a diagram showing the system of the prior art, in which thenumber of resresh cycles is ½ √{square root over (n)} whereas the numberof sense amplifiers to be simultaneously activated is 2√{square rootover (n)} for a capacity of n bits;

FIG. 111 is a diagram showing one embodiment of the DRAM having thestructure of ¼·n words×4 bits according to the present invention;

FIG. 112 is a diagram showing the DRAM having the structure of ¼·nwords×4 bits of the prior art;

FIG. 113 is a diagram showing the structure of the nibble mode accordingto the present invention;

FIG. 114 is a diagram showing the structure of the nibble mode of theprior art;

FIG. 115(a) is a diagram comparing the address systems having thestructure of n words×1 bit of the prior art and according to theembodiment of the present invention;

FIG. 115(b) is a diagram comparing the address systems having thestructure of ¼·n words×4 bits of the prior art and according to theembodiment of the present invention;

FIG. 116(a) is a diagram showing the package contour and pinarrangements of the DRAM of 16 Mbits according to one embodiment of thepresent invention;

FIG. 116(b) is a diagram showing the package contour and pinarrangements of the DRAM of 4 Mbits according to one embodiment of thepresent invention;

FIG. 117(a) is a diagram showing the memory cell structure using thestacked capacitor STC adopted in one embodiment of the presentinvention;

FIG. 117(b) is a diagram showing the memory cell structure using thehigh-speed plate capacitor HSPC adopted in one embodiment of the presentinvention;

FIG. 118(a) is a diagram combining the refresh system using the presentinvention and the in-chip voltage converter;

FIG. 118(b) is a diagram showing the method for the in-chip voltage VCLto charge the bit lines through the sense amplifier;

FIG. 119 is a diagram showing the DRAM of the prior art using no voltageconverter;

FIGS. 120(a) and 120(b) are a diagram and a time chart showing the DRAMhaving a √{square root over (n)} refresh×4 structure for introducing theA_(i−1) address from the I/O pins;

FIG. 121(a) is a diagram showing another embodiment of the presentinvention;

FIG. 121(b) is a circuit diagram for changes to the ½√{square root over(n)} refresh cycle only for the column address signal CAS and the beforerow address signal RAS;

FIG. 122 is a circuit diagram for forming a sense amplifier activationsignal from the refresh cycle switching address signals A_(xi−1) andA_(xi−1) shown in FIG. 121(b);

FIG. 123(a) is a diagram showing a peak current increase preventingcircuit of single-phase drive type;

FIG. 123(b) is a diagram showing a peak current increase preventingcircuit of two-phase drive type;

FIGS. 124(a) and 124(b) are diagrams showing another embodiment of thepresent invention;

FIG. 125 is a chip layout of the DRAM using one embodiment of thepresent invention;

FIG. 126(a) is a diagram showing the case in which the number of therefresh cycles is 2√{square root over (n)} in the present embodiment;

FIG. 126(b) is a diagram showing the case in which the number of therefresh cycles is 4√{square root over (n)} in the present embodiment;

FIG. 127 is a circuit diagram showing one embodiment of the voltage dropcircuit of the dynamic RAM to which is applied the present invention;and

FIG. 128 is a block diagram showing one embodiment of the dynamic RAMcontaining the voltage drop circuit of FIG. 127.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a fundamental layout showing one embodiment of a dynamic RAMto which is applied the present invention.

The RAM will have its operating speed dropped, as the various wiringlengths of control signals or memory array drive signals are elongatedas a result of an increase in the chip size accompanying an increase inthe memory capacity. In order to prevent this speed drop, the presentdevice has made the following devices in the arrangement of memoryarrays composing the RAM and peripheral portions for address selections.

In FIG. 1, there is provided a cross area which is formed in the chipwith a longitudinal center portion and a transverse center portion. Thiscross area is mainly arranged with peripheral circuits, and the fourareas divided by the cross area are arranged with memory arrays.

The cross area is divided, as shown in FIG. 1, into areas A to D.Specifically, the area A is located at the lefthand side of thetransverse center of the chip, and the area A is located at therighthand side of the transverse center of the chip. The area C islocated at the upper side of the longitudinal center of the chip, andthe area D is located at the lower side of the longitudinal center ofthe chip. Moreover, an area E is located at the center of the chip, inwhich the vertical center portion and the longitudinal center portion ofthe chip intersect each other.

In the memory chip of this embodiment, the four areas divided by thecross areas A to E are constructed of memory arrays. Each of these fourmemory arrays is made to have a memory capacity of about 4 Mbits,although not limitative, as will be especially limitative. Accordingly,the four memory arrays totally have a memory capacity as high as about16 Mbits.

Of the cross areas, the peripheral portions adjoining each of the memoryarrays is arranged with a decoder and a driver for selecting the memoryarrays. Specifically, the two memory arrays divided longitudinally bythe areas A and B are corresponded to by Y (Column) decoders (Ydec) andY select (Column select) drivers (YS drivers) the two memory arraysdivided transversely by the areas C and D are corresponded to by X (Row)decoders (Xdec) and word line drivers (WL drivers). As a result, thefour memory arrays are arranged by extending the word lines transverselyand the data lines (bit lines or digit lines) longitudinally. Since,however, one memory array has a capacity as large as about 4 Mbits, thenumber of the memory cells to be connected with one data line isimpractically enlarged. As a result, each memory array is composed ofsuch a plurality of memory mats as will be described hereinafter.

The remaining portions of the cross areas A to E are arranged with thefollowing major circuit blocks. The areas A and B are arranged withaddress buffers, address comparators (redundancy decoders), controlclock generators, data input buffers and so on. The areas C and D arearranged with common source switch circuits, sense amplifier controlsignal circuits, mat selection control circuits, main amplifiers and soon. And, the center area E is arranged with X-decoder and Y-decoderaddress signal generators, an internal voltage-drop power circuit and soon.

FIG. 2 is an overall layout showing one embodiment of the dynamic RAMaccording to the present invention. Specifically, the portioncorresponding to the aforementioned area A is arranged with Y-circuitsincluding a Y-address buffer, a Y-redundancy circuit and a Y-addressdriver (logical step), a test function circuit and a CAS control signalcircuit. Closer from the area A to the center, there are arranged aninternal voltage-drop VDL limiter circuit for transforming an externalpower voltage VCCE such as about 5 V to a voltage such as about 3.3 V tobe fed to a memory array, and a Y-address driver, an X-address driverand a mat selection driver DV1 to DV3.

The portion corresponding to the aforementioned area B is arranged withX-circuits including an X-address buffer, an X-redundancy circuit and anX-address driver (logical step), a RAS control signal circuit, a WEcontrol signal circuit and a data input buffer. Closer from the area Bto the center, there are arranged an internal voltage-drop VCC limitercircuit for transforming an external power voltage VCCE such as about 5V to a voltage such as about 3.3 V to be fed to a peripheral circuit,and a Y-address driver, an X-address driver and a mat selection driverDV1 to DV3.

If, as in the areas A and B, the address buffers, the redundancycircuits containing the address comparators corresponding to the addressbuffers, and CAS and RAS control signal circuits for generating controlclocks are concentrated in one position, a high integration can beachieved by distributing the clock generators and other circuits acrossthe wiring channel, namely, by sharing the wiring channels. At the sametime, the signals can be transmitted at the shortest equal distance sothat the operations can be speeded up.

The portion corresponding to the aforementioned area C is arranged withfour main amplifiers corresponding to totally eight memory mats arrangedsymmetrically with respect to the center axis of the area C, an internalboost voltage circuit VCHG, a substrate voltage generator VBBG, and fourmain amplifiers corresponding to the remaining totally eight memory matsarranged symmetrically with respect to the center axis of the area C.Thus, in this embodiment, one memory array is arranged with eight memorymats, and two memory arrays are arranged symmetrically with respect tothe area C so that totally sixteen memory mats are provided. Thanks tothis arrangement, the main amplifiers can have their number decreasedand their signal propagation distances shortened to speed up theoperations.

The portion corresponding to the aforementioned area D is arranged withfour main amplifiers corresponding to totally eight memory mats arrangedsymmetrically with respect to the center axis of the area D, four dataoutput buffers, and four main amplifiers corresponding to the remainingtotally eight memory mats arranged symmetrically with respect to thecenter axis of the area D. Thus, this embodiment is constructed of fourmemory arrays, as described above, so that the memory mats are thirtytwo in total number.

In this embodiment, the aforementioned longitudinal center area isarranged with bonding pads indicated by small square symbols, althoughnot especially limitative. The detail arrangement of these bonding padsis specifically shown in the layout of FIG. 3. In FIG. 3, the bondingpads indicated by solid squares are those for external power supply. Inorder to increase the input level margin, i.e., to decrease the powerimpedance, totally thirteen pads VSS for supplying the ground potentialof the circuit are arranged on one line. These pads VSS are connectedwith longitudinally extending ground potential leads which are formed bythe LOC technology. Of these pads VSS, the pad disposed in each of theareas C and D is used as a ground potential for preventing the floatingdue to the clearing of the word lines and the coupling of thenon-selected word lines of word drivers. Two pads disposed in each ofthe areas C and D are provided for the common source VSS of a senseamplifier to drop the wiring resistance of the common source there by tospeed up the operations. The area D is further arranged with two padsfor the data output buffer, and the area E is arranged with pads whichare operative to supply the ground potential to the X-address buffer andthe Y-address buffer and which correspond to the power generator.Moreover, one pad of each of the areas C and D and two pads of the areaE correspond to other peripheral circuits. As a result, the groundpotential of the circuit has a lower power impedance for the operationsof the internal circuit, and the VSS wires between five kinds ofinternal circuits thus divided are connected through low-pass filterscomposed of LOC lead frames and bonding wires, so that not only thegeneration of noises but also the propagations of the VSS noises betweenthe internal circuits can be minimized.

The pads corresponding to the external power VCCE such as about 5 V aretwo at the center corresponding to internal drop voltage generator VCClimiter and VDL limiter for the aforementioned voltage transformations,and one in a position corresponding to the data output buffer. Thesepads are also used for dropping the power impedance and for suppressingthe noise propagations of the voltages (VCC, VDL and VCCE) of theinternal circuits.

Address input pads A0 to A11 are arranged altogether at the center.These arrangements are intended, in proximity according to thearrangements of the X-address buffer and the Y-address buffer, tominimize the signal transmission distances thereby to speed up theoperations.

Control signal pads RAS, CAS, WE and OE are arranged close to theirrespectively corresponding circuits. Data output pads DQ1 to DQ4 aredisposed for data output buffers. A pad D is for data input of 1-bitstructure, and a pad Q is for data output of 1-bit structure.

In addition to the above-specified external pins, this embodiment isequipped with pads for bonding masters, monitoring and monitoring padcontrols. For these bonding masters, there are provided pads FP0 andFP1, the former of which is used to designate an SC (static column) modeand the latter of which is used to designate the write mask function ofan NB (nibble) mode×4-bit structure. The monitoring pads are designatedat VCC, VDL, VL, VBB, VCH and VPL. These pads are used for monitoringthe corresponding internal voltages VCC, VDL, VL, VBB, VCH and VPL. Theinternal voltage VCC is the peripheral circuit power voltage of about3.3 V; the voltage VDL is the power voltage of about 3.3 V to besupplied to a memory array, i.e., a sense amplifier; the voltage VCH isthe boost power voltage at a selection level for the word line boostedto about 5.3 V in response to the internal voltage VDL, i.e., forselecting a shared switch MOSFET; the voltage VBB is a substrate backbias voltage of −2 V; the voltage VPL is a plate voltage of a memorycell; and the voltage VL is a reference voltage of about 3.3 V for theVCC limiter and the VDL limiter. Pads VBT, VHT and VPLG are used forcontrolling the monitoring pads, as will become apparent from the laterdescription of the monitor voltage functions.

In this embodiment, the bonding pads are arranged in two rows. Moreover,the arrangement is made alternate with a pitch shift of one half. Inother words, the plural bonding pads are arranged zigzag. Thisarrangement can elongate the substantial distance between the pads. Inother words, a number of bonding pads can be arranged in high density ina relatively small area. The bonding pads have to make their pitchesrelatively large partly because they require a relatively large area tobe occupied thereby for the Wire bonding or the like and partly becauseit is necessary to provide an electrostatic breakdown preventingcircuit. Thus, the zigzag arrangement like this embodiment makes itpossible to arrange the numerous bonding pads in the relatively smallarea. On the contrary, the structure, in which the bonding pads arearranged in the longitudinal center portion of a long chip, makes itpossible to provide a number of pads like the aforementioned one.

FIG. 4 is a block diagram showing one embodiment for address assignmentsfor the memory arrays of the aforementioned structure.

The RAM of this embodiment has a memory capacity of about 16 Mbits, ashas been described hereinbefore. Moreover, the address signals take theaddress multiplex method, in which X-address signals and Y-addresssignals are fed in time-series in synchronism with the address strobesignals RAS and CAS. Of the address signals, the X-address signals arecomposed of 12 bits of X0 to X11, and the Y-address signals are alsocomposed of 12 bits of Y0 to Y11. As shown, the address signals X0 toX11 are true signals indicating the selection state, when the addresssignals fed from the outside are at a high level, and address signalsX0B to X11B are bar signals indicating the selection state, when theaddress signals fed from the outside are at a low level. Likewise, theaddress signals Y0 to Y11 are true signals indicating the selectionstate, when the address signals fed from the outside are at the highlevel, and address signals Y0B to Y11B are bar signals indicating theselection state, when the address signals fed from the outside are atthe low level.

The memory mat has a minimum unit of two regions SL and SR across thesense amplifier, and the corresponding X-decoder, word line driver andcolumn selector. The quartered memory array is arranged with eight-unitmemory mats. These unit memory mats are classified into eight kinds,i.e., MS0L and MS0R to MS3L and MS3R. Since each quartered memory arrayhas eight unit memory mats, the memory mats MS0L and MS0R to MS3L andMS3R are assigned to the four unit memory mats.

The X-decoder of the aforementioned unit memory mat is fed with theaddress signals of 8 bits of the address signals X0 to X7, the SL and SRsignals for designating the two regions across the sense amplifier, andthe signals MS0L/R to MS3L/R for designating the memory mat. One memorymat has 512 word lines. The unit memory mat takes the so-called “sharedsense amplifier type”, in which complementary data lines (e.g., bitlines or digit lines) are arranged at the righthand and lefthand sidesof the sense amplifier. Moreover, the address signals X8 and X8B areused as the righthand and lefthand address designating signals SL andSR. Thus, the X-decoder has a function to select one word line bydecoding the address signals of substantially 9 bits X0 to X8.

The address signals of 3 bits of the address signals X9 to X11 form themat selection signals MS1L/R. Specifically, the address signals X9 andX9B select the adjoining memory mats, as shown at MS0L and MSIL asrepresentatives in FIG. 4, and the address. signals X11 and X11B selectseither of two groups of righthand and lefthand memory blocks eachcomposed of two adjoining memory mats, as shown at MS0L and MS1L and atMS)R and MS1R as representatives in FIG. 4. Moreover, the addresssignals X10 and X10B are used to select either of the memory arrayswhich are divided by the areas of the longitudinal center portion of thesame Figure. By the aforementioned combination of the address signals of3 bits, the aforementioned eight address assignments MS0-3L/R aredesignated at the memory mats of each unit.

When the X-address signal is received in synchronism with the rowaddress strobe signal RAS, the selections of the X-line areaccomplished. At this time, one side of those two of the four memoryarrays, which are divided across the area of the longitudinal centerportion, is selected in response to the address signal X10 or X10B bythe aforementioned address assignments. In response to the addresssignal X11 or X11B, moreover, one of the memory mats having the letter Ror L added is selected, and one of the adjoining memory mats isdesignated in response to the address signal X9 or X9B. In the four ofthe totally thirty two memory mats, therefore, there is selected oneword line which is designated by the address signals (X0 to X8) of 9bits.

The Y-decoder corresponding to each memory array (composed of totallyeight memory mats) decodes the Y-address signals Y2 to Y9 to select thecomplementary data lines of the memory array. Specifically, theY-decoder decodes the address signals of 8 bits Y2 to Y9 to accomplishthe address selections of 1/256. However, the column selector selectsthe complementary data lines at a unit of 4 bits. Thus, one memory mathas a storage capacity of 512×256×4, and one memory array has eightmemory mats, so that the memory array has a total storage capacity of512×256×4×8=4,194,304, i.e., about 4 M bits. Since the DRAM is composedof four memory arrays, it has a storage capacity as high as about 16 Mbits.

Here, one memory block is constructed of totally eight memory mats—onegroup being composed of four memory mats MS0L to MS3L whereas the othergroup being composed of four memory mats MS0R to MS3R. For this memoryblock, there are provided four main amplifiers MA.

When the row address is thus decided, one of the eight memory mats MS0Lto MS3L and MS0R to MS3R composing the aforementioned one memory blockis selected in response to the address signals of 3 bits X10 and X10B,X11 and X11B, and X9 and X9B so that the aforementioned signals of 4bits are outputted in a manner to correspond to the four mainamplifiers.

One of the four main amplifiers AS0 to AS3 is selected in response tothe address signals Y0 and Y1 of the Y-address signals. In response tothe remaining address signals Y10 and Y11, moreover, one of the fourgroups of main amplifiers NA0 to NA3 is selected. Thus, one of thetotally sixteen main amplifiers is activated in response to the addresssignals of 4 bits Y0 and Y1, and Y10 and Y11 so that the read signal of1 bit is outputted through a data output circuit.

Incidentally, in the case of the memory access at the unit of 4 bits,although not especially limitative, the address signals Y10 and Y11 maybe ineffective to output the signals of those totally four mainamplifiers of the four groups, which are designated by the addresssignals Y0 and Y1, in parallel with one another. In the readingoperations in the nibble mode, moreover, although not especiallylimitative, the aforementioned main amplifiers may be addressed andadvanced stepwise in response to the address signals Y0 and Y1 or Y10and Y11 to output signals of 4 bits serially.

FIG. 7 is a schematic layout for specifically explaining theaforementioned power supply lines and the relations between theirassociated internal power circuit and the pads.

Reference numeral 1 designates the external power pad VCCE for supplyingthe power voltage to an internal drop power circuit (VCC) 3 through awiring layer. The internal drop power circuit (VCC) 3 receives thesupply of the power voltage VCCE such as about 5 V and generates theinternal voltage VCC such as about 3.3 V for the peripheral circuit. Theinternal voltage VCC is transversely fed through a wiring line 5 tooperate the address buffers or decoders. Moreover, the wiring line 5 isbranched at a generally center portion into two halves, which areextended vertically or longitudinally. These extensions correspond tothe aforementioned power supplies of the X-decoders and the mainamplifiers. Thus, the wiring lines 5 are branched not only vertically,as above, but also transversely in positions corresponding to theredundancy circuits.

Reference numeral 2 designates the external power pad VCCE for supplyingthe power voltage VCCE to an internal drop power circuit (VDL) 4 througha wiring layer. The internal drop power circuit (VDL) 4 receives thesupply of the power voltage VCCE such as about 5 V and generates theoperating voltage VDL such as about 3.3 V for the memory arrays (orsense amplifiers). The voltage VDL is distributed generally in the formof two squares sharing one side through wiring lines 6. Specifically,the wiring lines 6 are once extended transversely from the output pointof the internal drop power circuit (VDL) 4 and then arranged in arectangle enclosing the longitudinally extended wiring lines 5. Thus,the wiring lines 6 take the aforementioned form of the two squaressharing one side. Numeral 7 designates a power pad for the data outputbuffer and the guard ring, which is extended transversely in parallel toenclose the pad or main amplifier at the longitudinally center portion.The extensions of the power pad 7 at the two vertical ends enclose thewhole area of the chip, thus establishing the guard ring function.

FIG. 8 is a schematic layout for specifically explaining the groundlines of the aforementioned circuit and the relations between theassociated internal power circuit and the pads.

Reference numerals 11 located at the upper and lower ends of the centerportion of the chip designate ground potential supplying pads VSS forword clearing and word line latching operations. The pads 11 are onceextended transversely and then branched in positions corresponding tothe word drivers until they are extended vertically. The pads 11 arefurther extended transversely and further at the ends corresponding tothe word clearing portions until they are connected to each otherNumerals 12 designate ground potential pads for the common sources ofthe sense amplifiers and for supplying the ground potential to activatethe sense amplifiers. In this embodiment, the ground potential pads 12are arranged vertically symmetrically with respect to the transversecenter portion. At the upper side, the pads 12 are two in number and arerespectively extended transversely and then vertically in positionswhere there are provided the power switches MOSFETs for supplying theground potential to the sense amplifiers. Numeral 13 designates anelement for supplying the ground potential to the data output buffers.This element 13 is composed of two pads arranged to correspond to thefour data output buffers and wiring lines connecting the pads. Numeral14 designates a ground potential pad for the internal drop powercircuits VCC and VDL and the address buffer. The ground potential pad 14is connected with the wiring lines which are extended transversely tothe right and left. Numeral 15 designates a ground potential pad foranother circuit, i.e., for supplying the ground potential to theremaining circuits such as the aforementioned decoder circuits or mainamplifiers. Since the number and range of the circuits to be suppliedwith the ground potential are large and wide, the pads are as many asfour, and the wiring lines to be connected with the pads are extendedrelatively complicatedly in the transverse and longitudinal directions,as shown. In this embodiment, the ground lines are divided into one tofive kinds in accordance with the circuit functions and are commonlyconnected through the lead frame having the LOC structure. As a result,the noise leakage between the circuits having the ground lines dividedas above can be suppressed to enlarge the noise margin. Since the groundpotential is applied through the independent pads 14 and the relativelyshort wiring lines to the address buffers having severe restrictionsupon the noise margin, it is possible to retain a sufficient input noisemargin. This aims at effecting a substantial separation of a portionsuch as a sense amplifier, which is operative to generate relativelyhigh noises in the ground line, from the aforementioned circuit whichhas the severe restrictions upon the noises.

FIGS. 9(A) and 9(B) are a specific layout and a section showing an inputprotection circuit which is provided for the aforementioned bondingpads.

In this embodiment, as is apparent from the layout (A) and the section(B) of a portion of the former, the protection element is exemplified bya bipolar transistor of lateral type, i.e., N⁺-PWELL(Substrate)-N⁺,although not especially limitative. In this case, the emitter uses boththe voltages VCCE and VSS. When a high (positive/negative) voltage isapplied to the input, its potential is damped in this lateraltransistor. In this embodiment, as shown in the layout of FIG. 9(A), thepotential to be transmitted to the input gate is dropped by ahigh-resistance element made of polysilicon. The resistance value ofthis high-resistance element cannot be raised so high from thestandpoint of the transmission speed of the input signal but to anappropriate level such as 300 to 500 ohms in view of the signaltransmission function and the protection function.

The guard rings are formed around the NWELLs (N-type well regions) andmade of N⁺ to prevent any abnormal voltage at the input portion fromadversely affecting the peripheral circuits. These guard rings aresupplied with the voltage VCCE from the outside. In case the bondingpads are arranged in the center portion of the chip as in thisembodiment, the memory arrays or the peripheral circuits are more liableto be influenced by the surge voltage than the conventional case inwhich the bonding pads are arranged in the peripheral portion of thechip. Thus, the influences of the surge voltages through the substrateare reduced by surrounding the bonding pads by the guard rings as thewelled diffusion layers and by supplying the external power voltage VCCEto the guard rings.

Moreover, the aims at using the lateral type bi-polar transistors as inthis embodiment are as follows. Since the lateral type transistor canhave its area reduced, it is unnecessary to prevent the current frombeing concentrated by enlarging the opposed length (or base width) ofthe N⁺-type diffusion layers or the collectors or emitters thereby todrop the current value per unit length and to add any special processfor forming the transistor.

In FIG. 9, characters AL2 designate a second aluminum layer, andcharacters AL1 designate a first aluminum layer. Moreover, letters SiLdesignate a passivation open layer, and letters TC designate throughholes for connecting the second aluminum layer AL2 and the firstaluminum layer AL1.

FIG. 10 is a specific layout showing an input protection circuit to beprovided for the external power voltage VCCE pad.

When a high voltage is applied to the VCCE pad, its charges are releasedto the ground potential VSS by the lateral type bipolar transistor ofNWELL-PWELL (substrate)-NWELL. This protection element is disposed ineach of the upper and lower ends of the longitudinal center of the chip.As a result, the high voltage can be dropped at the inlet of the leadsrunning longitudinally at the chip center portion of the LOC structure.Thanks to this structure, the protection element is provided not in aone-to-one relation to the plural power pads but only in the paired padsin the vicinity of the inlet of the leads, so that the padscorresponding to the center portion of the leads can be kept away fromthe high voltage.

FIG. 11 is a layout showing the peripheral portion of the semiconductorchip, and FIG. 12 is a section showing a portion of FIG. 11 and a memorycell omitted from FIG. 11.

This embodiment adopts the structure, in which the peripheral circuitsand the bonding pads are arranged in the longitudinal and transversecenter portion of the chip, as has been described hereinbefore. As aresult, the chip is arranged with the memory arrays eve at itsperipheral portion and four corners. In this case, the four chip cornersmay possibly be cracked in the passivations by the stress of the resinof the package. In order to prevent this cracking, namely, to increasethe mechanical strength, the step of the memory array, as shown, isutilized to form an FG (i.e., the polysilicon gate electrode of the MOStransistor) and a WSi/PolySi (i.e., the polysilicide layer forming thecomplementary data lines). As shown in schematic section in FIG. 12,moreover, the first aluminum layer AL1 and the second aluminum layer AL2are superposed through an inter-layer insulating film. The stress comingfrom the resin is prevented from being applied directly to the memoryarray portion by forming such a gentle step in the corners of the chip.Moreover, the stress can be dispersed by elongating the corner portionsFG and WSi/PolySi.

As shown in the layout of FIG. 11 and in the section of FIG. 13,moreover, there is arranged in the outermost periphery of thesemiconductor chip a P⁺-type diffusion layer which is supplied with thesubstrate bias voltage VBB from the first aluminum layer AL1 and thesecond aluminum layer AL2. Inside of the P⁺-type diffusion layer, thereis arranged as the guard ring the NWELL which is formed at its centerwith an N⁺-type region which is supplied with the external power voltageVCCE from the first aluminum layer AL1 and the second aluminum layerAL2.

The aforementioned guard ring by the NWELL has a function to absorb theminority carriers which are generated from the P⁺-type diffusion layersupplied with the substrate bias voltage VBB when a voltage of about −2V generated by the substrate back bias voltage generator VBBG isabruptly changed by some cause. As a result, the minority carriersgenerated from the P⁺-type diffusion layer can be prevented frommigrating to the memory arrays and and from being coupled to theinformation charges stored in the memory capacitors of the memory cells,so that the information charges may be prevented from being reduced orbroken.

FIG. 5 is a block diagram showing the dynamic RAM according to thepresent invention while noting the control signals. FIG. 5 is drawn tocorrespond to the layout shown in FIG. 2.

The RAS control circuit is used to activate the X-address buffer inresponse to the signal RAS. The address signals are fed from theX-address buffers to the X-redundancy circuits. Here, the addresssignals are compared with the defective addresses stored, and it isdecided whether or not the switching should be made to the redundancycircuits. The decided results and the aforementioned address signals arefed to X-predecoders. Here, pre-decode signals Xi and AXn1 are generatedand fed through X-address drivers XiB and AXn1 corresponding to theindividual memory arrays, to the X-decoders corresponding to theaforementioned memory mats. In the same Figure, only one driver is shownas a representative by way of example.

On the other hand, the internal signals of the RAS system are fed to thecontrol circuits of the WE system and the CAS system. By deciding theinput order of the RAS signal, CAS signal and WE signal, for example,the automatic refresh mode (CBR) and the test mode (WCBR) arediscriminated.

In the test mode, the test circuit is activated to set the testfunctions in response to a specific address signal fed at that time.

Of the address signals taken into the X-address buffers, the addresssignal for instructing the selection of a memory mat is transmitted tothe mat selector MSiL/R, from which is selected any of the plural memorymats disposed in each memory array. Here, letters CS designate a commonsource switch MOSFET which is provided for a memory mat.

Like the address assignments shown in FIG. 4, the four main amplifiersMA correspond to four pairs of complementary data lines (of 4 bits)leading from the totally eight memory mats and arranged symmetricallywith respect to the main amplifiers. In response to the memory matselection signal MSiL/R, one of the eight memory mats is selected. Theseselections are accomplished by a unit mat control circuit UMC. In thesame Figure, four pairs of main amplifiers MA are shown as one group byway of example, and the remaining three groups of main amplifiers areshown in black boxes drawn by broken lines.

The mat selector MSiL/R generates the selection signals MS0L/R toMS3L/R. If the signal MS0L is generated, for example, the four memorymats corresponding to the signal MS0L shown in FIG. 4 are selected.These four memory mats MS0L are made to correspond to the aforementionedfour main amplifiers MA because they have input/output nodes of 4 bits.

The control circuit of the CAS system is used to generate a variety ofY-control signals in response to the signal CAS. The address signalstaken into the Y-address buffers in synchronism with the change of thesignal CAS to the low level are fed to the Y-redundancy circuits. Here,the address signals are compared with the defective addresses stored,and it is decided whether or not the switching should be made to theredundancy circuits. The decided results and the aforementioned addresssignals are fed to Y-predecoders. Here, pre-decode signals Yi and AYn1are generated and fed through Y-address drivers (at the final step) YiBand AYn1 corresponding to the individual four memory arrays, to theX-decoders corresponding to the aforementioned memory mats. In the sameFigure, only one driver YiB and AYn1B is shown as a representative byway of example.

On the other hand, the aforementioned CAS control circuit activates theadjoining test circuit when it decides the input sequence of the RAS andWE signals received there to decide the test mode.

The bonding pads to be fed with the address signals and the controlsignals are gathered and arranged at the center portion of the chip,although omitted from the same Figure. As a result, the distances fromthe individual pads to the corresponding circuits can be shortened to agenerally equal value. Thus, by taking the layout of this embodiment,the address signals and the control signals can be taken at a high speedto minimize the skew which is caused among the address signals ofmultiple bits.

As shown in the same Figure, moreover, the sense amplifier (SA) powersource VDL and the peripheral circuit power source VCC are also arrangedat the center portion of the chip. As a result, the circuits arranged atthe four corners of the chip can be supplied with the various voltagesthrough the wiring lines of equal and short distances. In each circuit,moreover, capacitors having a relatively large capacitance for voltagestabilizations, i.e., for dropping the power impedance are dispersed inthe circuit along the power wiring lines, although not shown.

FIG. 6 is a block diagram noting the operation sequence for thestructure of×1 bit. In FIG. 6, the circuit blocks are indicated mainlyby signal names, and the major circuits are indicated by circuit names.Thus, the signal paths showing the flows of the write/read signals areomitted from FIG. 6.

With reference to FIG. 6, the operations of the dynamic RAM will beschematically described in the following.

The address selecting operations of the row system are accomplished inthe following.

The address signal Ai (A0 to A11) and the address signals A9 to A11 andA8 are taken into the address buffers in synchronism with the rowaddress strobe signal RAS so that they are retained as the row internaladdress signal BXi, MSiL, MSIR, SL and SR. The address signal BXi takeninto the address buffers is inputted on one hand to the redundancycircuit to decide whether or not the memory access is to the defectiveaddress. On the other hand, the address signal BXi is fed to thepredecoder to generate a predecode signal AXNL, which is inputted to theX-decoder X-DEC provided for each memory mat. For the address signals A8to A11, there are provided another set of buffers MSIL, MSiR, SL and SR.In short, the address signals A0 to A11 have their loads made relativelyheavy because they are fed to the redundancy circuit and the predecoderso that they are inputted the numerous address comparators and gatecircuits in the redundancy circuit. Thanks to the provision of the matselecting address buffers MSiL, MSiR, SL and SR, according to thisembodiment, the operations are speeded up because of no influence of thedelay in the signals, which influence is caused by the input capacitiesof the redundancy circuit and the predecoder.

The X-decoder X-DEC is fed with an X-decoder precharge signal XDP and anX-decoder extraction signal XDG, which are generated from the matselection signals MSiL/R, SL and SR for controlling the operation timingof the X-decoder X-DEC. The X-decoder X-DEC decodes the aforementionedpre-decode signal AXNL in response to the timing signals SCP and SDG togenerate the selection signals of the word lines. If, at this time, theaccesss is to the defective address, a signal XRiB to be outputted fromthe redundancy circuit is generated to inhibit the selections of theword lines with the output of the X-decoder X-DEC and to select theredundant word line. For these word line selecting operations, theaforementioned boosted voltage VCH is used. Thus, the transmissions ofthe signal charges between the memory cells and the complementary datalines can be accomplished without any level loss without any relation tothe threshold voltage owned by the address selecting MOSFET having itsgate coupled to the redundant word line.

The aforementioned mat selection signals MSiL/R generate a complementarydata line precharge signal PCB. Since the memory mat to be selected isdecided by the mat selection signals MSiL/R, the precharging operationsare released (or ended) at the complementary data lines of the selectedmat. Then, the selection signals SL/SR are generated to designate thelefthand region SL or righthand region SR of the memory mat designatedby the address signal A8. From the signals SL/SR and the mat selectionsignals MSiL/R, there is generated a selection signal SHR forcontrolling the switch MOSFET for selecting the region SL or SR to becoupled to the sense amplifier. The selection signal SHR to be used hereis exemplified by the aforementioned boosted voltage VCH. As a result,the signals are transmitted without any level loss between the senseamplifiers and the complementary data lines selected.

The sense amplifiers are activated, if the conditions for controlsignals PN1 and PP1 of the power switch MOSFET prepared from the RASsignal, the selection signals of the word lines and the mat selectionsignals MSiL/R are satisfied. At this time, the sense amplifiers areactivated by the voltage VDL which has been dropped in the inside, ashas been described hereinbefore. At this time, the two-stepamplifications are accomplished, although not shown, because of the dropin the peak current according to the operations of the sense amplifiers.At the first step, more specifically, the switch MOSFET for a relativelysmall current is turned ON to activate the sense amplifiers. At thesecond step for the relatively large amplified output, the switch MOSFETfor a relatively large current is turned ON to effect the high-speedamplifications.

A signal RG determines the timing at which the Y-switch MOSFET is turnedON. After a sufficient signal is attained on the complementary datalines, the signal RG is generated to control the timing for selecting alater-described column system.

Signals RN and RF decide the normal read mode and the refresh mode. Ifthe signal CAS is changed from the high to low levels before the signalRAS is changed from the high to low levels, the signal RF is generatedto cause the refresh mode (i.e., CAS before RAS refresh). In this case,the address selections of the column system to be accomplishedthereafter are omitted by a signal CE.

If the signal CAS is changed from the high to low levels when the signalRAS is at the low level, the normal mode signal RN is generated. Inresponse to this, the signal CE for the read/write controls isgenerated. An address signal BYi taken in the Y-address buffer is fed tothe Y-redundancy circuit and the predecoder to generate a pre-decodesignal AYNL. A signal AC1B controls the operations of the mainamplifiers and the Y-decoders and is generated as the address signalsare changed at the break or at the high level of the signal CE.

A signal YiB is generated in the absence of the relief address in theredundancy circuit, whereas a signal YRiB is generated in the presenceof the relief address.

The Y-decoder Y-DEC decodes the pre-decode signal AYNL, in the absenceof the defect relief, to generate a Y-(column) selection signal andinvalidates the address selection corresponding to the pre-decode signalAYNL, in the presence of the defect relief, to generate a reliefY-(column) selection signal.

A write signal W2 is prepared from the signal WE. From the signal CAS,there is prepared a signal C2 which is used for controlling the RAS/CASlogic, the read/write discriminaiton, and the setup and holdingcharacteristics. A signal W3B is a one-shot pulse for read-modify-writeoperations and early-write operations to generate internal write pulses.

A signal WYP is used for controls from the data input buffers toinput/output lines I/O, and a signal WYPB takes charge of controls fromthe input/output lines I/O to the complementary data lines. A signal DLdetermines a data setup/hold time when a write signal Din is taken intothe data input buffers. A write data DDi taken into the data inputbuffers is transmitted in response to the signal WYP.

The write signal of the input/output line I/O is transmitted to thecomplementary bit (or data) line selected by the Y-decoder circuit Y-DECand is written in one memory cell which is coupled to the complementarybit line to have its word lines selected.

The signal YP is an operation control signal for the Y-decoder system,and a signal RYP is an operation control signal for the main amplifier.The former signal YP is generated in the write operation, too, becauseit controls the Y-decoder Y-DEC.

From the signal RYP, activation signals MA and RMA are prepared toactivate the main amplifier. A signal DS controls the data output timingof the main amplifier.

From the mutual input timing relations of the signals RAS, CAS and WE,there are respectively prepared test mode signals RN and RF, signals WNand WF, and signals CR and LF. The signals RN and RF and the signals WNand WF control CBR (CAS before RAS refresh) and WCBR (WE, CAS beforeRAS). The signals CR and LF control the testing circuits, e.g.,sets/resets the address signal Ai in the aforementioned WCBR. Theaddress signal AFi taken into the testing circuit is converted into FMiBfor determining test modes to generate various test signals.

As the power circuit, there is prepared from the voltage VCCE such asabout 5 V, which is fed from the external terminal, a drop voltage VCCsuch as about 3.3 V for a peripheral circuit, from which is furtherprepared a bootstrap voltage VCH such as about 5.2 V for determining theselection level of the word lines. Moreover, the voltage VCC is used toprepare the substrate back bias voltage VBB at about −2 V. From thevoltage VCCE fed from the outside, moreover, there are independentlyprepared the drop voltage VDL such as about 3.3 V for the memory array(or sense amplifiers) and the drop voltage VST which is fed when in thestandby.

From the schematic operations described above, the plural memory matseach composed of the memory arrays contain the X-decoders for selectingthe word lines. These X-decoders are fed, as shown in the block diagramof FIG. 5, through the final driver stages with the mat selectionsignals MSiL/R prepared by the mat selection circuits MSiL/R and thepre-decode outputs AXNL and XiB prepared by the predecoder. Theindividual circuits arranged at the center portion are corresponded toby the address inputting bonding pads, the control signal RAS, theaddress buffers and the redundancy circuits. As a result, the wiringlength for transmitting the address signals can be shortened to speed upthe operations. For example, in the layout in which the bonding pads arearranged on the two shorter sides of a rectangular chip such as the DRAMof the prior art to distribute the address terminals and the controlterminals accordingly, the signal transmission length are elongatedaccording to the size of the chip. In other words, there are mixedlonger and shorter distances from the bonding pads to the inputterminals of the address buffers. The distances from the address buffersto the address decoders become accordingly longer and shorter accordingto the positions of the address buffers. Thus, this layout system aimingat taking a large storage capacity as large as about 16 Mbits has itsoperating speed dropped in proportion to the size of its chip partlybecause the operating speed is determined by the longest one of thesignal paths for handling the signal lines and partly because it isnecessary to take the timing margin.

In the DRAM of this embodiment, on the contrary, the address inputtingbonding pads and the control inputting bonding pads are concentrated atthe center portion, as has been described hereinbefore, and the addressbuffers and the control circuits are arranged in proximity. Since, inthis structure, the signal lines extend radially from the center portionof the chip, the signal propagation distances can be shortened to aboutone half of the size of the chip. The wiring resistance and wiringcapacity get larger in proportion to the wiring length. Hence, thesignal propagation delay time is delayed on principle in proportion tothe square of the signal propagation distance. It follows that thesubstantial reduction of the signal propagation distance to one half ofthe size of the chip causes the reduction of the signal propagationdelay time to one quarter.

The embodiment takes a structure in which only the memory mat at theunit selected by the mat selection signal MSiL/R is activated. On thebasis of the mat selection signal MSiL/R, there are generated for eachmemory mat the signals SHR and PCB necessary for the address selectingoperations of the mat and the sense amplifier activation signal.According to this structure, a timing margin for the aforementionedsignals SHR and PCB and the sense amplifier activation pulses need notbe taken between the memory mat arranged at a relatively short distancefrom the mat selection circuit arranged at the center and the memory matarranged at a longer distance. In other words, the memory mat thusactivated starts its operation, when it is fed with the aforementionedmat selection signals MSiL/R, to generate a variety of signals for theaddress selections by the timing system optimized in the subsequent unitmat.

According to this structure, the mat selection circuit to be arranged atthe center portion of the chip is sufficient, if it feeds eight matselection signals to thirty two mats in the foregoing embodiment, sothat it can lighten the signal load and reduce the number of signallines. As a result, the delay of the selection signals to be transmittedto each mat can be reduced. Moreover, the memory mat thus selectedoperates at the timing optimized for each mat and needs no timing marginbetween the mats so that a high-speed memory access can be accomplished.

As in the address assignments of the memory mats shown in FIG. 4, on theother hand, the two axially symmetric memory mats, e.g., MS0L and MS1Lor MS2L and MS3L constitute one sub-block. Four sub-blocks are providedfor one memory array. In this structure, only one of the aforementionedtwo axially symmetric memory mats is activated. Thus, one controlcircuit can be used commonly for the two memory mats.

In the aforementioned sub-block composed of two memory mats, the memorymats, e.g., MS0L, MS1L, MS2L and MS3L, which are in axially symmetricrelation between the memory arrays separated by the longitudinal centerarea, may be used as one memory block for one control circuit. In thiscase, too, only one of the memory mats MS0L, MS1L, MS2L and MS3L isactivated so that one control circuit can be commonly used In this case,eight memory blocks are made for all the memory arrays.

The control circuit is effective, if it generates only one of thevarious signals such as the aforementioned ones for the prechargingoperations of the complementary data lines, the activations of the senseamplifiers, the shared sense amplifier controls, the activations of theX-decoders, the activations of the word drivers, the activations of theY-decoders, the selections of the common input/output lines I/O, and theselections and activations of the main amplifiers. The control circuitcan be more effective if it can generate all of the various signals.

In case the memory array is to be constructed as the block of the unitmats, the number of operable mats can be easily changed merely bychanging the mat selection circuit, i.e., the mat selection logic. Thus,the kind development (to a lower power) can be easily accomplished.

Moreover, the X-decoders or Y-decoders for selecting the word lines ordata lines may be disposed either adjacent to the memory mat of the unitor commonly for the plural unit mats. In this embodiment, the X-decoderis disposed for each memory mat, whereas the Y-decoder is disposed foreach memory array so that it may be shared among the eight memory matsto provide an efficient layout.

FIG. 14 is a fundamental layout showing another embodiment of thedynamic RAM according to the present invention.

In this embodiment, like the aforementioned embodiment of FIG. 1, eachof the four memory arrays, which are divided by the cross area composedof the longitudinal center portion and the transverse center portion, isequipped with a Y-decoder. In this structure, the Y-decoder is arrangedat the center of each memory array so that the column selection linescan be shortened. This makes it possible to speed up the selectingoperations of the Y-system. In accordance with this structure, thepre-decode signals of the Y-system are fed to the individual Y-decodersthrough wiring channels disposed in the longitudinal center portion.Here, what are disposed at the sides adjacent to the longitudinal centerportion are X-decoders similar to the aforementioned ones.

In this structure, too, similar speed-up can be achieved by arranging atthe center portion of the aforementioned chip the bonding pads, theinput circuit such as the corresponding address buffers, the memorymats, the sub-blocks or the memory block selector.

FIG. 15 is a fundamental layout showing another embodiment of thedynamic RAM according to the present invention.

In this embodiment, like the aforementioned embodiment of FIG. 1, thefour memory arrays, which are divided by the cross area composed of thelongitudinal center portion and the transverse center portion, areequipped with X-decoders at their respective center portions. Accordingto this structure, the lengths of the word lines at each unit memory matcan be halved to accordingly lighten the loads upon the word lines sothat the word line selecting operations can be speeded up. In accordingto this structure, the pre-decode signals of the X-system are fed to theX-decoders corresponding to the individual memory mats through thewiring channels formed in the X-decoders. Here, what are disposed at thesides adjacent to the transverse center portion are the Y-decoders likethe aforementioned ones.

In this structure, too, similar speed-up can be achieved by arranging atthe center portion of the aforementioned chip the bonding pads, theinput circuit such as the corresponding address buffers, the memorymats, the sub-blocks or the memory block selector.

FIG. 16 is a fundamental layout showing another embodiment of thedynamic RAM according to the present invention.

In this embodiment, like the foregoing embodiment of FIG. 1, the fourmemory arrays, which are divided by the cross area composed of thelongitudinal center portion and the transverse portion of the chip, areequipped with X-decoders and Y-decoders in the longitudinal andtransverse directions. According to this structure, the lengths of theword lines and the column selection lines can be halved to lighten theloads accordingly to speed up the word line selections and the columnline selections. In this structure, one of the four memory areas dividedby the X- and Y-decoders may be selected and equipped at its centerportion with control circuits for generating a variety of signals forthe precharging operations of the complementary data lines, theactivations of the sense amplifiers, the controls of the shared senseamplifiers, the activations of the X-decoders, the activations of theword drivers, the activations of the Y-decoders, the selections of thecommon input/output lines I/O, and the selections and activations of themain amplifiers.

In this structure, too, similar speed-up can be achieved by arranging atthe center portion of the aforementioned chip the bonding pads, theinput circuit such as the corresponding address buffers, the memorymats, the sub-blocks or the memory block selector. Incidentally, theX-decoders and the Y-decoders may be interchanged in the foregoingembodiments of FIGS. 14 to 16.

Whatever of the modifications of the aforementioned fundamental layoutsmight be adopted, the memory arrays are quartered by the cross regionsof the longitudinal and transverse center portions of the chip so thatthey may be formed with the peripheral circuits and the bonding pads.Especially in the structure which is arranged at its center with theaddress pads, the address buffers, the predecoders or the final-stagedriver for feeding the pre-decode signals to the individual decoders,the propagation paths of the signals for the memory access are radiallyextended longitudinally and transversely at the shortest equal distanceto the four corners. As a result, the aforementioned high-speedoperations can be accomplished.

The chip is also arranged generally at its center portion with theinternal power source such as the drop voltage generator for generatingthe operating voltage VDL of the memory arrays (or sense amplifiers) orthe operating voltage VCC of the peripheral circuits. According to thisstructure, the wiring length for the power supply can also be shortenedlike the foregoing embodiment of FIG. 7. As a result, the powerimpedance can be suppressed at a low level to speed up the operations ofthe circuit and drop the noises.

FIG. 17 presents layouts showing the fundamental structures of anotherembodiment of the memory mat and another embodiment of the memory blockconstructed by composing the fundamental structures.

FIG. 17(A) shows the fundamental structure of the memory mat. In FIG.17(A): letter S designates a sense amplifier; letter M a memory cellarray; letter W a word line driver (including an X-decoder); and letterC a control circuit. In the embodiment of FIG. 17(A), the senseamplifier S is disposed at the lefthand side of the memory cell array M.Therefore, the memory mat of this embodiment does not adopt the sharedsense amplifier system unlike the foregoing embodiment.

FIG. 17(B) shows the structure in which the memory cell arrays M arearranged symmetrically with respect to the sense amplifier S of theaforementioned memory mats to construct the sub-block. In this case, thesense amplifier S may be used, by the shared sense amplifier system,selectively for either of the righthand and lefthand memory cell arraysM, or two sense amplifiers S may be arranged adjacent to each other in amanner to correspond to the individual memory cell arrays. A pluralityof these sub-blocks are combined to construct the aforementioned memoryarray. If, in this structure, the righthand and lefthand memory cellarrays are selected, the control circuit C can be shared.

FIG. 17(C) shows one memory block, which is constructed by combining thememory mats of FIG. 17(A) such the word line drivers W, the memory cellarrays M and the sense amplifiers S are arranged verticallysymmetrically to have the control circuit C located at the center of thesub-block of FIG. 17(B). In this case, each of the paired verticallysymmetrical sub-blocks may be composed of two memory arrays. By makingsuch an address assignment that one of the four quartered memory cellarrays M (or unit memory mats) may be selected, the sense amplifier S iscoupled selectively to the righthand and lefthand memory cell arraysthrough the switch MOSFET by the shared sense amplifier system, and theword line drivers W may be shared between the upper and lower memorycell arrays. According to this structure, the control circuit can beshared for the four memory mats. In this case, however, the Y-signalcircuit is eliminated because there is no Y-decoder in the mats orblocks.

FIG. 18 presents layouts showing the fundamental structures of anotherembodiment of the memory mat and another embodiment of the memory blockconstructed by composing the fundamental structures.

FIG. 18(A) shows the fundamental structure of the memory mat. In theembodiment of FIG. 18(A), the control circuit C is disposed adjacent tothe sense amplifier S. The-word line drivers W are disposed at the twovertical sides of the memory cell array M. The word line drivers W areused to bring one word line into its selected/unselected state so as toeffect the high-speed selecting operations of the word line. Thisstructure may be replaced by another structure in which the word line ofthe memory cell arrays M is vertically bisected so that the two wordlines may be selected by the aforementioned two word line drivers W. Inthis case, the word lines can be selected at a high speed by shorteningthe word lines. Moreover, the word lines may be alternately selected bythe two vertical word line drivers. According to this structure, thepitch of the word lines to be selected can be twiced for the verticallydivided word line drivers. In other words, the word line driversrequiring a relatively large area of occupation are vertically dividedso that they can drive the word lines which are arranged at a smallerpitch. The memory mats of this embodiment do not adopt the shared senseamplifier system like the foregoing embodiment.

FIG. 18(B) shows a sub-block which is constructed by arranging thememory cell arrays M and the corresponding sense amplifiers Ssymmetrically with respect to the control circuit C of theaforementioned memory mat. In this case, the control circuit C isshared. There may be adopted the shared sense amplifier system, in whichthe control circuits C are vertically distributed and the senseamplifier S may be shared and used selectively for the two memory cellarrays.

FIG. 18(C) shows one memory block which is constructed by arranging thememory cell arrays M, the sense amplifiers S and the control circuit Cvertically symmetrically with respect to the word line drivers W of theaforementioned sub-block. In this case, those of the quartered memorycell arrays (or unit memory mats), which constitute the sub-block, maybe constructed into two memory arrays. By making such an addressassignment that one memory cell array M of the aforementioned memoryblock may be selected, the control circuit can be shared for the memoryblock among the four memory mats. In this case, however, the signalcircuit of the Y-system is eliminated because no decoder of the Y-systemis present in the mat or block.

FIG. 19 presents layouts showing the fundamental structures of anotherembodiment of the memory mat and another embodiment of the memory blockconstructed by composing the fundamental structures.

FIG. 19(A) shows the fundamental structure of the memory mat. In theembodiment of FIG. 19(A), the sense amplifiers S are disposed at therighthand and lefthand sides of the memory cell array M. As a result,the complementary data line (or bit line) of the memory cell array M isdivided at the center. Thus, the number of the memory cells of thecomplementary data lines to be coupled to the inputs of the senseamplifiers can be halved to reduce the parasitic capacitance thereby tolighten the load and to increase the amount of signals read out from thememory cells so that the sense amplifiers S can be speeded up. Thisstructure may be replaced by another in which the sense amplifiers S areconnected with the two ends of the complementary data lines to amplifythe signals read out from the two ends of the complementary data lines.According to this structure, the currents of the sense amplifiers can bedispersed to speed up the operations and to drop the noises.

On the other hand, the sense amplifiers may be distributed to the rightand left alternately of the pair of the complementary data lines. Inthis case, it is possible to loosen the pitch of the sense amplifiers.In other words, by the aforementioned distribution of the senseamplifiers, one of these sense amplifiers can be formed in an areacorresponding to two pairs of complementary data lines so that the pitchof the complementary data lines can be made more tight. Below the memorycell arrays M, there is disposed the word line driver W which isenclosed by the control circuit C.

FIG. 19(B) shows a sub-clock which is constructed by arranging twomemory mats symmetrically with respect to one sense amplifier S of theaforementioned memory mat. In this case, the control circuit is shared.In case the word line of the righthand and lefthand memory cell arraysis selected only alternately, there may be taken the modified sharedsense amplifier system in which the central sense amplifiers S areshared and selectively used for the two memory cell arrays if, in thiscase, the sense amplifier at the center is used for auxiliaryamplifications, its input/output are connected directly with one end ofthe complementary data lines of one memory cell array, and the other endis coupled without any difficulty to the input/output of the senseamplifier through the switch MOSFET.

FIG. 19(C) shows a memory block which is constructed by arranging fourmemory mats vertically symmetrically with respect to the control circuitC of the aforementioned sub-block. In this case, those of the quarteredmemory cell arrays M (or unit memory mats), which constitute thesub-block, may constitute each of the two memory arrays. By making suchan address assignment that one memory cell array M of the aforementionedmemory block may be selected, the control circuit can be shared for thememory block composed of the four memory mats. In this case, however,the signal circuit of the Y-system is eliminated because there is nodecoder of the Y-system in the mat or block.

FIG. 20 presents layouts showing the fundamental structures of anotherembodiment of the memory mat and another embodiment of the memory blockconstructed by composing the fundamental structures.

FIG. 20(A) shows the fundamental structure of the memory mat. In theembodiment of FIG. 20(A), the sense amplifiers S are disposed at therighthand and lefthand sides of the memory cell array M, and the wordline drivers W are disposed above and below the memory cell array M. Asa result, the complementary data line (or bit line) of the memory cellarray M is divided at the center. Thus, the number of the memory cellsof the complementary data lines to be coupled to the inputs of the senseamplifiers can be halved to reduce the parasitic capacitance thereby tolighten the load and to increase the amount of signals read out from thememory cells so that the sense amplifiers S can be speeded up. Thisstructure may be replaced by another in which the sense amplifiers S areconnected with the two ends of the complementary data lines to amplifythe signals read out from the two ends of the complementary data lines.According to this structure, the currents of the sense amplifiers can bedispersed to speed up the operations and to drop the noises. For thehigh integration like the aforementioned embodiment, moreover, the senseamplifiers may be alternately arranged at the two ends of thecomplementary data lines.

The word line drivers W are used to bring one word line into itsselected/unselected state so as to effect the high-speed selectingoperations of the word line. This structure may be replaced by anotherstructure in which the word line of the memory cell arrays M isvertically bisected so that the two word lines may be selected by theaforementioned two word line drivers W. In this case, the word lines canbe selected at a high speed by shortening the word lines. Like theforegoing embodiment, moreover, the word line drivers may be arrangedalternately at the two ends of the word lines to effect the highly densearrangement of the word lines.

The control circuit C is arranged to enclose the word line driver belowthe memory cell array M and the lefthand sense amplifier.

FIG. 20(B) shows a sub-clock which is constructed by arranging twomemory mats symmetrically with respect to the lefthand control circuit Cof the aforementioned memory mat. In this case, the control circuit isshared. In case the word line of the righthand and lefthand memory cellarrays is selected only alternately, there may be taken the modifiedshared sense amplifier system in which the central sense amplifiers Sare shared and selectively used for the two memory cell arrays. If, inthis case, the sense amplifier at the center is used for auxiliaryamplifications, its input/output are connected directly with one end ofthe complementary data lines of one memory cell array, and the other endis coupled without any difficulty to the input/output of the senseamplifier through the switch MOSFET.

FIG. 20(C) shows a memory block which is constructed by arranging fourmemory mats vertically symmetrically with respect to the control circuitC below the aforementioned sub-block. In this case, those of thequartered memory cell arrays M (or unit memory mats), which constitutethe sub-block, may constitute each of the two memory arrays. By makingsuch an address assignment that one memory cell array M of theaforementioned memory block may be selected, the control circuit can beshared for the memory block composed of the four memory mats. In thiscase, however, the signal circuit of the Y-system is eliminated becausethere is no decoder of the Y-system in the mat or block.

FIG. 21 presents layouts showing the fundamental structures of anotherembodiment of the sub-block and another embodiment of the memory blockconstructed by composing the fundamental structures.

In FIG. 21(A), the sub-blocks shown in FIG. 17(B), each which iscomposed of the memory arrays M arranged at the righthand and lefthandsides of the sense amplifier S, the word line drivers W arranged belowthe memory cell arrays M and the control circuit C arranged therebelow,are either arranged symmetrically or juxtaposed, and a Y-decoder to beused commonly for the memory cell arrays M is disposed at the righthandside.

In FIG. 21(B), the memory block shown in FIG. 18(C) is equipped withcommon X-decoders. In this embodiment, the letter W designates merelythe word line drivers which have no decoding function. If, in thisembodiment, only one of the four memory cell arrays M selects the wordlines, the word line drivers may be shared by the two memory cellarrays.

Even the foregoing structures of the memory mat, the sub-block and thememory block, as shown in FIGS. 17 to 21, can activate the unit memorymat in response to a suitable mat selection signal. In response to thismat selection signal, there are generated for each memory mat thesignals SHR and PC necessary for selecting the address of the mat andthe sense amplifier activation signals. According to this structure, notiming margin is required for the signals SHR and PC and the senseamplifier activation signals between the memory mat, which is arrangedat a relatively short distance from the mat selector arranged at thecenter, and the memory mat which is arranged at a relatively longdistance. In other words, the memory mat to be activated starts itsoperation, at the instant when it is fed with the mat selection signals,and the various signals for the unit mat activations are then generatedby the timing system which is optimized in the unit mat. Thus, the matselector arranged at the center of the chip may supply the selectionsignals for activating any of the aforementioned mats so that the signalloads can be lightened to reduce the number and delay of the signals tobe transmitted to each mat. Moreover, the memory mat thus selected canoperated at the timing optimized therefor, but no timing margin isrequired between the mats so that the memory access can be accomplishedat a high speed.

FIG. 22 is a top plan view showing the SOJ (Small Outline J-bendpackage) lead frame to be used in the. DRAM according to the presentinvention.

The packaged DRAM chip is indicated by double-dotted lines in FIG. 22. Apair of leads extended transversely through the center of the chip areused for supplying the ground potential VSS and the power voltage VCCE.With this arrangement of the leads across the center of the chip, theleads are bonded to the plural power pads VSS and VCCE. Moreover, thepower terminals are composed of the aforementioned two terminals VCCEand VSS, and the ground potential VSS and the power voltage VCCE are fedto the plural portions of the chip through the wiring material having alow resistance such as the lead frame so that the power impedance to befed with those potentials can be suppressed at a low value. As a result,it is possible to suppress the noises which are caused in the powerlines by the operation current of the circuit.

On the other hand, the leads for exchanging the signals are formed tohave their connection ends extended from the top and bottom to thecenter of the chip, as shown. Thus, the connections to the addresssignal terminals and the control terminals, which are collected at thecenter of the chip, can be efficiently. accomplished.

FIGS. 23(A) to 23(C) show an example of the connection between the leadframe and the semiconductor chip.

In the example of FIG. 23(A), a lead frame 22 and the surface of a chip23 are connected through a film 24 by an adhesive A 26 and an adhesive B27. Moreover, the lead frame 22 has its terminal connected through agold wire 25 to the bonding pad of the chip 23.

In the example of FIG. 23(B), the lead frame 22 is connected by anadhesive C 29 to an insulator 28 which is formed over the surface of thechip 23. The terminal of the lead frame is connected through the goldwire 25 to the bonding pad of the chip 23.

In the example of FIG. 23(C), the lead frame 22 is covered with a moldresin 21 except its surface portion to be connected for the bonding andis connected by an adhesive D 30 to the surface of the chip 23. Theterminal of the lead frame is connected through the gold wire 25 to thebonding pad of the chip 23.

In case such lead frame is used, it can be so arranged on the surface ofthe semiconductor chip as to form a part of wiring lines. As a result,the bonding pads can be arranged without any problem at the centerportion of the chip and can be connected to the leads.

FIG. 24(A) is. an exterior view showing the DRAM of the LOC (Lead OnChip) structure using the aforementioned lead frame, and FIG. 24(B) is aperspective view showing the inside of the DRAM.

In these Figures: reference numeral 31 designates a mold resin, numeral32 a an external terminal (or lead frame); and numeral 33 a chip. Thischip 33 is coupled through an insulating film 34 to the lower side ofthe lead by means of the aforementioned adhesive. In the inside, eachlead has its leading end connected through a gold wire 35 to a bondingpad 38 of the chip 33. Numeral 36 designates a bus bar lead which isused for supplying the aforementioned voltages VCCE and VSS. Numeral 37designates a suspension lead, and numeral 39 designates an index.

FIG. 25(A) is a diagram showing the pin arrangement of the externalterminals. The aforementioned dynamic RAM of 16 Mbits is packed in thepackage having twenty eight bins, although not especially limitative.FIG. 25(B) is a side elevation showing the side at which the pins arearrayed, and FIG. 25(C) is a section taken along the side at which thepins are not arrayed.

FIG. 26 is a diagram showing the arrangements of the pins of structuresof ×1 bit and ×4 bits in case the ZIP (Zigzag In-line Package) is usedin the dynamic RAM according to the present invention. In FIG. 26,letters NC designate empty pins, and the arrows appearing in the DRAM ofthe structure of ×4 bits indicate the same signal pins as those of thestructure of ×1 bit.

FIG. 27 is a diagram showing the arrangements of the pins of thestructures of ×1 bit and ×4 bits in case the SOJ type package is used inthe dynamic RAM according to the present invention. In FIG. 27, lettersNC designate empty pins, and the arrows appearing in the DRAM of thestructure of ×4 bits indicate the same signal pins as those of thestructure of ×1 bit.

In case the lead frame of the aforementioned LOC structure is used, thebus bar lead extending the longitudinal direction of the chip is used asthe ground potential VSS of the circuit, and the ground potentialsupplying pads are provided at the side of the DRAM chip in a manner tocorrespond to the operation units of the chip so that the groundpotential may be supplied from the plural portions. According to thisstructure, the ground potential is applied directly to the circuit ofeach operation unit from the lead frame of the low impedance so that ahigh level margin can be taken at the ground potential side. Another busbar lead extending the longitudinal direction of the chip is used forthe external voltage VCCE, and the power pad is provided for the circuitrequiring the external voltage VCCE such as the data output buffer orthe internal drop voltage generator VCC or VDL. As a result, the powerimpedance can be dropped to reduce the power noises resulting from theinternal operations. Especially the output buffer for generating theoutput signal is used to feed a high drive current for driving arelatively large load. Therefore, the special power pads VCCE and VSSare arranged for and in the vicinity of the aforementioned outputbuffer, thus making it possible to reduce the noises and to prevent thenoises, if any, from adversely affecting another circuit.

The dynamic RAM according to the present invention will be specificallydescribed in the following with reference to specific circuit diagramsand their operation waveforms.

In the following specific circuit diagrams, signals designated atletters terminated by B such as a signal WKB are bar signals using theirlow level as the active level.

FIG. 28 presents circuit diagrams showing portions of one embodiment ofthe control circuit of the RAS system. FIG. 70 is a timing chart showingone embodiment of the individual signals of the RAS system.

The RAS (Row Address Strobe) signal is fed to the input circuit havingthe CMOS inverter structure. This CMOS inverter circuit for the inputbuffer is constructed of P-channel and N-channel MOSFETs having anabsolute value of threshold voltage of about 0.5 V, although notespecially limitative. By equalizing their conductance ratio, moreover,the CMOS inverter circuit is made to have a logic threshold voltage ofabout 1.6 V. The power voltage VCC for the peripheral circuit of theDRAM of this embodiment is set at 3.3 V which is about two times as highas the logic threshold voltage of 1.6 V. These voltage values hold ateach input buffer which is made receptive of another control signal CASor WE, an address signal or a write data. The aforementioned logicthreshold voltage corresponds to the signal TTL level.

In the DRAM aiming at increasing the capacity as in this embodiment, theelement is made fine. In the circuit dislike the dispersion of theelement constant like the MOSFET composing the internal invertercircuit, therefore, the portion having flat characteristics in thechannel length Lg—the threshold voltage Vth is used. As a result, thechannel length Lg is relatively enlarged to raise the threshold voltageVth to a relatively high level so that the operating speed is dropped incase the DRAM is operated at the relatively low voltage VCC.

Therefore, the MOSFET composing the initial-stage inverter circuit ofthe input buffer required to have the high speed is set, although notespecially limitative, to have the aforementioned low threshold voltageby making its channel impurity concentration lower than that of theMOSFET composing the inverter circuit used as the internal circuit. TheMOSFET having such low threshold voltage is also used as the inputinitial-stage circuit for other control signals or address signals.Moreover, the MOSFET having the low threshold voltage from the viewpointof the operating speed or the level drop is used as the output stageMOSFET of the output buffer in the DRAM of the CMOS structure like thatof this embodiment, the initial-stage MOSFET of the main amplifier, thepull-up MOSFET of the input/output lines I/O, the short MOSFET of thecomplementary data lines, or the MOSFET of diode mode used in the chargepump circuit. Incidentally, the method of attaining the aforementionedlow threshold voltage can take a variety of modes of embodiments inaddition to the aforementioned one in which the impurity concentrationof the channel is changed by the ion implantation.

The DRAM is rendered operative, if the signal RAS is set at the lowlevel, and inoperative if the signal RAS is set at the high level.

The RAS signal having passed through the inverter circuit acting as theaforementioned input buffer is introduced through a NAND gate circuit,which uses the signal WKB as its gate control signal, into the latchcircuit which is composed of two NAND gate circuits having their inputsand outputs connected crossly.

The aforementioned signal WKB is raised to the high level when the levelof the substrate back bias voltage VBB is shallow. As a result, theoutput of the inverter circuit takes the low level to fix the output ofthe NAND gate circuit at the high level so that the reception of thesignal RAS is inhibited. When the substrate back bias voltage is notsufficient, the operations of the internal circuit cannot be warrantedto inhibit the RAM access. Moreover, the output of the NAND gate circuitis positively fed back to the gate of the P-channel MOSFET which isdisposed at the input of the NAND gate circuit. Between the P-channelMOSFET and the operating voltage VCC, there is connected in series theP-channel MOSFET which has its gate steadily fed with the groundpotential to act as a resistance element. As a result, once the signalRAS is taken into the aforementioned gate circuit, it is made reluctantto be inverted, by shifting the logic threshold voltage to the lowlevel.

If the substrate back bias voltage VBB is at the desired deep level, thesignal WKB is at the low level. Then, the NAND gate circuit is opened sothat the RAS signal having passed through the input buffer is taken intothe latch circuit The signal RE is a rewrite warrant signal to hold theinternal RAS signal when it is at the high level.

A signal R1 having passed through the aforementioned latch circuit isused to control the X-address buffers and input buffers for the matselections, CAS, WE or Din. In other words, the individual circuits areactivated in response to the high level of the signal R1. A signal R1Bis inverted from the signal R1.

From this signal R1, a delay signal R1D is formed by an inverter circuitof tandem connection (which will be shortly referred to as an “invertercircuit array”), and a signal R2 is formed by an inverter circuit and aflip-flop circuit. The signals R1 and R1D control the later-describedX-address buffer, i.e., determines the set-up/hold of the X-addresssignal.

The signal R2 is used to set/rest the word lines. Moreover, the resettiming of the word lines is delayed for compensating the write level.

From the signal R2, a signal FUS is formed by the flip-flop circuit, theinverter circuit and the NAND gate circuit. The signal FUS is used toset the initial value of a later-described redundancy circuit. Thesignal FUS is a one-shot pulse having a constant pulse width from thesignal R2 to feed an electric current to a defective address storingfuse for a constant period so that the level may be held in the latchcircuit as the fuse is cut or not. As a result, the defective addressmemory circuit is initialized. Thanks to the use of such one-shot pulse,no steady DC current is fed to the uncut fuse so that the powerconsumption is reduced.

From the signal R2, a signal R3 is formed by the inverter circuit arrayand the flip-flop circuit. This signal R3 is used to control thecomplementary data line system (e.g., the sense amplifier SA, thepre-charge PC or the shared sense SHR) and the redundancy decoderprecharge RDP. The reset timing is delayed to make a sufficient delayfrom the reset (R2) of the word lines thereby to reset the complementarydata lines.

From the aforementioned signals R1 and R3, a signal RDP is formed by theNAND gate circuit and the inverter circuit.

FIG. 29 presents circuit diagrams showing other portions of oneembodiment of the control circuit of the RAS system.

A signal WM is used to monitor the set timing of the word lines therebyto control the operations of the complementary data lines (or senseamplifier). Therefore, the signal WM is formed from signals XE and XRE0Bto XRE3B, which are formed by the later-described redundancy circuit.Otherwise the remedy address, the signals XRE0B to XRE3B are at the highlevel so that the signal WM is formed from the signal XE. In the remedyaddress, on the other hand, the signal WM is formed by setting thesignal XE and one of the signals XRE0B to XRE38 at the low level.

A signal P0 is formed from the aforementioned signals WM and R3. SignalsPN1 and PP1 are formed by delaying the signal P0 to determine theamplification timing at the first step. These signals PN1 and PP1 areused to form either delay signals, which are formed to have relativelylarge delay times by a multiplexer and a flip-flop circuit, or signalsPN2 and PP2 which are formed to have relatively small delay times by themultiplexer and three inverter circuit arrays. These signals PN2 and PP2are used to determine the second-stage amplification timing of the senseamplifiers. The aforementioned multiplexer is switched in the test modeto make variable the peak current of the sense amplifiers.

FIG. 30 presents circuit diagrams showing another portion of oneembodiment of the RAS control circuit.

The aforementioned signal PN2 is delayed to form a signal RG by a delaycircuit composed of a flip-flop circuit and an inverter circuit. Thesignal RG determines the timing at which the Y-(column) switch is to beturned ON. When a sufficient signal is attained on the complementarydata line by the amplification of the sense amplifier, the Y-(column)switch is opened to output the signal to the input/output line I/O.

The signal RG is delayed to form a signal RE by a flip-flop circuit. Thesignal RG is a rewrite warrant signal and is used when the RAS signal istimed out. In the dynamic memory cell for selecting the memory cell bythe address selections of the row system, more specifically, theinformation charges of the information storage capacitor are once to bebroken by the selections, but the information storage charges areretrieved by the rewriting operation, in which the amplified output ofthe sense amplifier is received as it is. As a result, even if the RASsignal is raised to the high level before the aforementioned rewrite,the time period for this rewrite is retained by the high level of theaforementioned signal RE.

FIG. 31 is a circuit diagram showing one embodiment of a unit circuitconstituting the X-address buffer.

A NAND gate circuit made receptive of both an address signal AI fed froman external terminal and a signal R1 constitutes an input buffer. Inother words, the NAND gate has its gate opened, when the signal Ri takesthe high level, to take thereinto the address signal fed from theexternal terminal AI. In the input buffer having this gate function,too, the logic threshold voltage is set at the aforementioned level ofabout 1.6 V, and the operation voltage VCC is set at a twice level of3.3 V, as has been described hereinbefore. As a result, the logicthreshold voltage is set at a middle point of the operation voltage VCCso that the operation voltage can be efficiently used to enlarge theinput level margin.

A tri-state output circuit having its output high-impedance statecontrolled by a signal XLB is used as an input gate circuit for takingthe aforementioned address signal AI thereinto. A similar tri-stateoutput circuit to be controlled by a signal RLB is used as an input gatecircuit for taking a refresh address signal ARI thereinto. The addresssignal taken selectively through the two input gate circuits istransmitted to the input of a CMOS inverter circuit. Between the inputof this CMOS, there is connected a similar tri-state output circuitwhich is to be controlled by a signal XRLB, thus constituting an addresslatch circuit.

From the output of this address latch circuit, there are formed internaladdress signals BXI and BXIB through an inverter circuit and an NANDgate circuit.

From signals RID and CI, there are formed control signals XRLB, XLB andRLB for controlling the aforementioned tri-state output circuit.

Here, letter I designates one of the numerical values 0 to 11. In otherwords, the circuit appearing in the same Figure is a unit circuitcorresponding to each of the address signals A0 to A11. The unitcircuits corresponding to the address signals A0 to A11 have theiroutputs fed to the redundancy circuits of the X-system so that they areused as address signals to be referred to with the defective addressstored. For the address signals A8 to A11, moreover, there are providedthe following address buffer circuits for forming the memory matselecting signals

FIG. 32 is a circuit diagram showing one embodiment of the addressbuffer circuits corresponding to the address signals A9 and A10.

The address input circuit made receptive of the address signal fed fromthe external terminal, the input circuit of the refresh address signal,and the latch circuit shared by the former two circuits will not bedescribed because they are similar to that of FIG. 31. From the addresssignal taken into the aforementioned latch circuit, mat selectionsignals MS0B to MS3B are formed by an inverter circuit and a NAND gatecircuit. From the signals R3, RD1 and C1 of the row system, moreover,there are formed the control signals of input gates which constitute theaforementioned latch circuit.

FIG. 33 is a circuit diagram showing one embodiment of the addressbuffer circuit corresponding to the address signal A11.

The address input circuit made receptive of the address signal fed fromthe external terminal, the input circuit of the refresh address signal,and the latch circuit shared by the former two circuits will not bedescribed because they are similar to that of FIG. 31. From the addresssignal taken into the aforementioned latch circuit, signals BX11LB andBX11RB are formed by an inverter circuit and a NAND gate circuit. Thesesignals BX11LB and BX11RB select the righthand and lefthand sides of themats to operate. The signals BX11LB and BX11RB are outputted through theCMOS transmission gate circuit which is composed of an N-channel MOSFETand a P-channel MOSFET. The CMOS transmission gate circuit is switchedby the signal RC. The transmission gate circuit has its output sideequipped with the reset MOSFET for receiving the aforementioned signalRC.

Mat selection signal MSLIL and MSIR are formed from the aforementionedsignals BX11LB and BX11RB and the signal MSIB. Here, the letter Idesignates 0 to 3 so that the mat selection signals are composed ofeight kinds. From the signals R3, Rd1 and C1 of the row system,moreover, there are formed control signals XRLB, XLB and RLB of inputgates which constitute the aforementioned latch circuit.

In the normal mode, the signal RC is at the low level. As a result, theselection signals BX11LB and BX11RB of the righthand and lefthand matscorresponding to the address signals A11 and AR11 are formed through theaforementioned transmission gate circuit. In the test mode, on thecontrary, the signal RC is set at the high level. As a result, theaforementioned gate circuit is turned OFF so that both the signalsBX11LB and BX11RB are set at the low level by the reset MOSFET. Thismeans that the righthand and lefthand mats MSIL and MSIR aresimultaneously brought into the selected state. As a result, the refreshcycle in the test mode takes 2,084 cycles, which is half as high as4,096 cycles in the normal mode for the low level of the signal RC.Thus, in this embodiment, the refresh cycle can be switched.

FIG. 34 is a circuit diagram showing one embodiment of the addressbuffer circuit corresponding to the address signal A8.

The address input circuit made receptive of the address signal fed fromthe external terminal, the input circuit of the refresh address signal,and the latch circuit shared by the former two circuits will not bedescribed because they are similar to that of FIG. 31. From the addresssignal taken into the aforementioned latch circuit signals SLB and SRBare formed by an inverter circuit and a NAND gate circuit. These signalsSLB and SRB are used to generate the righthand and lefthand selectionsignals SL and SR in the mat selected: From the signals R3, RD1 and C1of the row system, moreover, there are formed the control signals XRLB,XLB and RLB of the gates which constitute the aforementioned latchcircuit.

The aforementioned address signals A0 to A11 are transmitted to thegates of a number of MOSFETs such as the address comparators in thepredecoder or redundancy circuit. As a result, the address buffer drivesa large capacitative load so that the signal of the internal addresssignal is relatively retarded. Thus, the mat selections to beaccomplished prior to the word line selections can be conducted at ahigh speed to achieve the high speed of the access time by providing themat selecting address buffers for the address signals A8 to A11.

FIG. 35 presents circuit diagrams showing portions of one embodiment ofthe predecoder of the row system.

Signals AXNLD and AXNLU are used for controlling the X-decoders, andupper and lower mats are selected by address signals BX10 and BX10B.

Signals AXIH and AXIHB accomplish controls (corresponding to the remedyof the defect in the sense amplifiers or the Y-(column) selection lines)of the redundancy decoders of the Y-system. Here, letter I designates anumeral of 8 to 11. The aforementioned signals AXIH and AXIHB are formedby setting/resetting the latch circuit composed of a pair of NAND gatecircuits in response to the signals BXIB and BXI. A signal AX10Hcontrols the upper and lower mats of the Y-decoders and the signals AYNLand YIB. A signal AXIH latches one cycle period of RAS for controllingthe Y-decoders.

FIG. 36 is a circuit diagram showing one embodiment of the redundancycircuit of the X-system. FIG. 72 is a timing chart showing thecorresponding operations.

The fundamental concepts of the redundancy circuit of this embodimentwill be described in the following.

The righthand and lefthand memory areas of each memory mats areindividually equipped with four redundancy word lines. According to onedefect remedy method of the DRAM of the prior art, a redundancy deocderis provided in one-to-one relation for each redundancy word line. Thelarge storage capacity provided by the numerous memory mats as in thisembodiment will increase the number of redundancy decoders.

In another defect remedy method of the DRAM of the prior art, a fuse isprovided for the enable of the redundancy decoder and the addresssignals X0 to X7. Since the redundancy word lines are simultaneouslyselected in the blocks of 2 16 which can be designated, their efficiencyis dropped, and the probability of presence of a defect in theredundancy word lines is increased to drop the defect remedyingefficiency.

Therefore, fuses are added to the address signals X8 to X11 so that theredundancy word line may be selected by only one of the sixteen blocks.In other words, the switching to the redundancy word line isaccomplished by only the block (or mat) in which a effective word lineis present. This operation is made possible by signals XR0DB to XR3DB(BX10) or XR0UB to XR3UB (BX10B) and the mat selection signals (MSiL/Rand XL/SR).

If the X-address direction is divided into sixteen by the four bits ofthe address X8 to X11, each block has four redundancy word lines so thatthe redundancy decoders to be disposed can be 4×16=64 at the most. Thismakes it possible to set the redundancy decoders at an arbitrary number(desirably a multiple of 4) from the minimum 4 to the maximum 64. In thepresent embodiment, the number of 12 is so selected from 4 to 64 thatthe remedy efficiency may be the best (to give the maximum yield). Theremedy efficiency of this defect remedying method can be substantiallyequalized to that of the case in which twelve redundancy word lines (orredundancy decoders) are disposed in another defect remedying method ofthe prior art. In other words, the number of redundancy word lines canbe reduced one third while leaving the number of redundancy decodersunchanged.

In FIG. 36, the fuse FUSE is made of a polysilicon layer and isselectively cut by the exposure to a laser beam in a manner tocorrespond to a defective address to be stored, although not especiallylimitative.

The fuse FUSE is initialized through the MOSFET, which is turned ON by asignal FUS of one shot pulse, and fixed at the ground potential by theMOSFET which is turned ON by the high output level of the invertercircuit if the fuse FUSE is cut. Otherwise, the input of the invertercircuit is fixed at the high level.

A signal RDP implies that the defect remedy is not accomplished if thefuse FUSE located at the upper side of the same Figure is not cut, and asignal XRDJB then takes the low level. Here, letter J designates one ofthe numerical values of 0 to 11, which correspond to the number oftwelve of the redundancy decoders. The fuse FUSE is cut for a defectremedy, and the signal XRDJB is raised to the high level in response tothe signal RDP.

In the same Figure, the upper fuse is for the enabling operation, andlower fuse is for storing a defective address. For the defect remedy,the enable fuse is cut. A signal XRDJ takes the high level when theaddress programmed in the redundancy decoder J is identical to the inputaddress X0 to X11. In the same Figure, the MOSFETs having their sourcesfed with signals XND0J to XND2J are of the N-channel type. A signalXRDJB takes the high level, when in the precharge, and the low level ifany of the input address signals X0 to X11 is different by only 1 bitfrom the address programmed in the redundancy decoder J, namely, if thedefect remedy address is not selected. The signal XRDJ takes the lowlevel when in the precharge and is left at the low level in case theremedy address is not selected.

When in a non-remedy state, the enabling fuse is not cut. As a result,the signal XRDJB is fixed at the low level, and the signal XRDJ is fixedat the low level. Signals A, B6 and B7 are used for testing theredundancy word lines In the test mode, a signal STB is dropped to the.low level. As a result, the redundancy decoders of J=0, 3, 6 and 9 arebrought into remedy state, and the address fuse is equivalently cut bythe combination (0, 0), (1, 0), (0, 1) and (1, 1) of X6 and X7, whichare caused to correspond to the four redundancy word lines XR0 to XR3,respectively, so that the redundancy word lines may be selected. At thistime, the address comparators of I=8 to 11 are brought into coincidentstates irrespective of the input addresses so that the redundancy wordlines are selected by all the sixteen blocks. Thus, it is possible toavoid the trouble that the redundancy word lines cannot be tested otherthan by one of the sixteen blocks.

In this embodiment, all the redundancy word lines are not necessarilyused, but it is more frequent not to use all of them. Noting this,according to this embodiment, the redundancy decoders are used commonlyfor selecting the redundancy word lines which are provided for theplural memory mats.

In this embodiment, there are provided two address comparators, as willbe reasoned in the following. In the redundancy decoder of the priorart, the coincidence is decided by the single address comparator, andthe selection pass of the ordinary word lines is then stopped inresponse to the decision. This system requires one-step logic forinhibiting the ordinary word line selection pass and the timing marginfor preventing the racing. In this embodiment, therefore, there areprovided two address comparators for detecting the coincidence andincoincidence. The redundancy word lines are selected, when thecoincidence is detected, and the ordinary word lines are selected if theincoincidence is detected. As a result, the one-step logic is reduced,and the timing relations of causing the racing of the prior art areeliminated so that the selections of the word lines can be speeded up.

FIGS. 37 and 38 are circuit diagrams showing decoder circuits forselecting the word lines and the redundancy lines.

In the circuit of FIG. 37, a signal XE is timing signal for selectingthe word lines when in the normal operations. When the aforementionedenabling fuse is cut, all the signals XRD0B to XRD11B take the low levelif word lines other than the defective one are accessed to. When any ofthe redundancy decoders of J=0 to 11 does not cut the enabling fuse,namely, when not in the remedy, signals BX0 to BX0B are dropped to thelow level to raise the signal XE to the high level. From the signals BX0and BX1, there are then formed pre-decode signals XKDB and XKUB (BX10and BX10B). Signals WCKDB and WCKUB are corresponding signals forclearing of the word lines (at their remote ends).

In the circuit of FIG. 38, a signal XRELB is a signal for selecting thefour redundancy word lines which are formed by dividing the twelveredundancy decoders into three sets. From this signal XRELB and thesignals BX10 and BX10B, there are formed redundancy word line selectionsignals XRLDB and XRLUB and redundancy word line clearing signals WCRLDBand WCRLUB in a manner to correspond to the upper and lower mats.

FIG. 39 presents circuit diagrams showing one embodiment of the timinggenerator for activating the sense amplifiers.

The ground potential is given by an N-channel MOSFET which is turned ONin response to a signal formed from the mat selection signal MSI and thesignal R3 by a timing signal PN1 for the first-step amplification. Theground potential is also given by an N-channel MOSFET which is turned ONin response to a signal formed by a timing signal PN2 for thesecond-step amplification. The operation voltage VDL is given by aP-channel MOSFET which is turned ON in response to a signal formed fromthe mat selection signal MSI and the signal R3 by a timing signal PP1for the first-step amplification. The operation voltage VDL is alsogiven by a P-channel MOSFET which is turned ON in response to a signalformed by a timing signal PP2 for the second-step amplification.

Incidentally, the power lines are so shared that the aforementionedN-channel MOSFET or P-channel MOSFET may not be erroneously turned ON bypower noises in case they are to be turned OFF when the ground potential(of the N-channel side) and the operation voltage VDL (of the P-channelside) of at least the final inverter of the circuit for controlling thegates of the N-channel MOSFET and the P-channel MOSFET are fed to thesense amplifiers, although not shown.

The N-channel MOSFET and P-channel MOSFET, which are turned ON at thefirst step, are caused to feed a relatively small current by havingtheir conductances reduced to a relatively small value. The N-channelMOSFET and P-channel MOSFET, which are turned ON at the second step, arecaused to feed a relatively large current by having their conductanceraised to a relatively large value.

The aforementioned mat selection signal MSI (I: 0L/0R to 3L/3R)activates the sense amplifiers of the four of the thirty two mats.

FIGS. 40 and 41 are circuit diagrams showing one embodiment of thecontrol circuit to be provided for the memory mats.

The circuit of FIG. 40 forms the following signals from mat selectionsignals MSIL/R, the signals SL and SR and the row timing signals R1 andR2. Here will be described the closed signal in one of theaforementioned thirty two mats. Therefore, the signals other than thesignals MSIL/R have their suffixes omitted. From the aforementionedsignals, there are formed X-decoder precharge signals XDPL/R, X-decoderextraction signals SDGLB/RB and a complementary data line pre-chargesignal PCB. On the other hand, word line driving signals WPHL/R and asignal MSH have their levels changed by a NOR gate circuit of latch typewhich is operated by the aforementioned bootstrap voltage VCH. Thesesignals at the changed high level are outputted through the invertercircuit which is operated by the aforementioned bootstrap voltage VCH.In the memory mat of this embodiment, therefore, the word lines to beselected are changed at a stroke from the non-selection level at the lowlevel to the boosted selection level. As a result, the word lineselections can be speeded up more than the structure of the prior art,in which the bootstrap voltage is established by using the word-lineselection signals and by combining their delayed signals.

The circuits of FIG. 41 are decoders and drivers for forming the wordlines WL and redundancy word lines RWL which are selected by theaforementioned pre-decode signal, the X-decoder precharge signals SCPL/Rand the X-decoder extraction signals SDGLB/RB.

The word line driver uses the aforementioned boosted voltage VCH as itsoperating voltage so that it raises the selected word lines linearlyfrom the ground potential VSS of the low level to the boosted voltageVCH.

Shared line drive signals SHL/R formed by the selection signals MSH, SLand SR are also controlled by the aforementioned boosted voltage VCH.Therefore, signals can be transmitted without any level loss by thethreshold voltage of the switching MOSFET between the sense amplifiersand the selected complementary data lines.

FIG. 42 is a circuit diagram showing one embodiment of the memory cellarray.

The memory cell is composed of an information storing capacitor and anaddress selecting MOSFET. This address selecting MOSFET has its drainconnected with one of a pair of complementary data lines arranged inparallel. The gate of the address selecting MOSFET is connected with theword lines. The other end (or plate) of the information storingcapacitor is supplied with the plate voltage.

In the same Figure, there are shown by way of example the pairedcomplementary data lines, the four word lines WL0 to WL3 and theredundancy word lines RWL0 to RWL3.

The coupling due to the overlap between the word lines and the pairedcomplementary data lines can be offset by the later-describeddifferential sense amplifiers because it appears on the complementarydata lines in the common mode. Incidentally, the complementary datalines are crossed at a constant interval and interchanged. Thus, theinfluences of the coupling of the complementary data lines can beeliminated.

With the remote ends of the aforementioned word lines, there areconnected word line clearing switch MOSFETs which are fed with theaforementioned clearing signals WCL0 to WCL3 and RWCL0 to RWCL3.

The complementary data lines are coupled to the input/output nodes ofthe sense amplifiers through the switch MOSFETs made receptive of ashared line drive signal SHL. One sense amplifier, as representativelyshown by way of example, takes the structure in which the input andoutput of the CMOS inverter circuit composed of P-channel and N-channelMOSFETs are crossly connected.

Incidentally in this embodiment, it should be noted that the senseamplifier means either the unit circuit, as described above, or thememory mat unit in which the unit circuits have a shared source.

In the sense amplifier, the P-channel MOSFETs have their common sourcePP fed with the operation voltage VDL through the power switch composedof the P-channel MOSFETs, and the N-channel MOSFETs have their commonsource PN fed with the ground potential VSS through the power switchcomposed of the N-channel MOSFETs, thus starting the amplifications ofthe sense amplifiers.

In this embodiment, there are provided column switch MOSFETs which areconnected with four pairs of input/output lines IO0 and IO0B to IO3 andIO3B each for the unit of four pairs of complementary data lines. Thus,a Y-(column) selection line YS is connected commonly with the gates ofthe four pairs of column switch MOSFETs. Accordingly, the redundancydata lines are composed of four pairs, and four sets of selectionsignals YSR0 to YSR3 are also given, although not shown.

FIG. 43 is a circuit diagram showing one embodiment of the refreshcounter circuit.

This CBR counter circuit accomplishes its counting operation to form arefresh address signal ARJ, when in the refresh mode, by using a signalRFDB corresponding to the RAS signal as its clock. A signal CAI is acarry input signal, and a signal CAJ is a carry-out signal. These twentyunit circuits are connected in tandem to generate refresh addresssignals AR0 to AR11 corresponding to the address signals A0 to A11. Inthis embodiment, the refresh operations of 4,096 bit scans areaccomplished.

FIG. 44 presents circuit diagrams showing portions of one embodiment ofthe control circuit of the CAS system. On the other hand, FIG. 75 is atiming chart showing one embodiment of the address selecting operationsof the CAS system.

The CAS (Column Address Strobe) signal is fed to the input circuitconstructed of a CMOS inverter circuit. This CMOS inverter circuit forthe input buffer is caused to have a logic threshold voltage such asabout 1.6 V like before. The operation voltage VCC is set at 3.3 V twiceas high as the aforementioned logic threshold voltage of 1.6 V andcorresponds to the signal at the TTL level. The operations of thecircuit of the Y-system are started when the signal CAS is dropped tothe low level.

The CAS signal having passed through the inverter circuit acting as theaforementioned input buffer is used in a circuit like that for theaforementioned RAS signal. However, the signal corresponding to thesignal WKB of the RAS circuit is omitted, and the power voltage VCC ofthe circuit is steadily supplied.

Signals C1 and C2 are formed from the signal CAS. The signal C1 is usedfor controlling the nibble counter, signals DoE, W3B and W5B and thesignal CE. The signal C2B is used for controlling the signal WYP, andthe signal C2 is used for controlling the signals W3B, YL and DL. Thesignal ACIB is formed from the signal CE so that the signals YP and RYPare formed.

The signal ACIB is a signal for controlling the operations of the mainamplifiers and the Y-decoders and is generated in response to the signalCE. A oneing operations in response to the signal AC1B. The signal YP isa signal for controlling the operations of the Y-decoders and isgenerated even when in the write operations. The signal RYP is a signalfor controlling the operations of the main amplifiers.

FIG. 45 is a circuit diagram showing one embodiment of the unit circuitconstituting the Y-address buffer.

A NAND gate circuit made receptive of both the address signal AI fedfrom an external terminal and the signal R1 constitutes an input buffer.Specifically, the NAND gate circuit has its gate opened, when the signalR1 takes the high level, to take thereinto the address signal fed fromthe external terminal AI. The signal R1 is used to reduce the electriccurrent in the standby state. In this standby state in which the signalR1 takes the low level, the input circuit is kept away from respondingto the signal of the address terminal AI. In the input buffer havingsuch gate function, too, the logic threshold voltage is set at about 1.6V, as has been described hereinbefore, and the operation voltage VCC isset at a twice level of 3.3 V. Thus, the logic threshold voltage is setat the middle point of the operation voltage so that the operationvoltage can be efficiently used to enlarge input level margin.

Tri-state output circuit, in which the output high-impedance state iscontrolled by the signal YL, is used as the input gate circuit fortaking the aforementioned address signal AI thereinto. A similartri-state output circuit to be controlled by the taken signal YL of theaddress signal constitutes a positive feedback loop between the inputand output of the CMOS inverter circuit, which is made receptive of theaddress signal through the aforementioned input gate circuit, toaccomplish the address latching operations. The output of this addresslatch circuit forms internal address signals BYI and BYIB through aninverter circuit.

The aforementioned internal address signals BYI and BYIB and the signalCE form together a signal ACIB.

The circuit for generating the signal YL is shown in FIG. 54, and theY-address buffer has four operation modes in accordance with thegeneration modes of the signal YL. The first mode is a normal mode, inwhich the signal YL changes in response to the CAS signal to make thestatic column operation possible. The second mode is a nibble mode, inwhich the signal YL is formed in response to the first CAS signal tohold the address signal taken. The third mode is a CBR mode, in whichthe CAS signal is reset and then dropped to the low level to generatethe signal YL so that the address signal is taken in. The fourth mode isa WCBR mode, in which the address signal accepted as effective betweenthe signals R1 and YL is taken in as the signal for designating the testmode.

FIGS. 46 to 49 are circuit diagrams showing one embodiment of theY-redundancy circuit and the predecoder circuit for remedying thedefects of the data lines, the column selection lines (which may besimply called the YS lines) or the sense amplifiers. The fundamentalconcepts of the redundancy circuits of the Y-system in this embodimentare similar to those of the aforementioned X-redundancy circuit.

Specifically, there are sixteen blocks which are divided by X8 to X11. Adefective data line of one block is remedied by the redundancy datalines. Thus, the address signals AX8H and AX8HB to AX11 and AX11B areinputted to the address comparator.

For four pairs of input/output lines I/O, four pairs of complementarydata lines are selected in one column selection line. As a result, theremedy is accomplished for the four pairs of complementary data lines.Therefore, the addresses Y0 and Y1 are restricted so that theircorresponding fuses are not provided. There is provided none of thefuses which correspond to the structure of ×4 bits and the addresses Y10and Y11 to be restricted in the nibble mode. As a result, fourredundancy YS lines simultaneously come out of one block. In thepractical layout, one block is divided in the direction of word linesinto four quarters (Y10 and Y11), which are longitudinally distributedand arranged in the chip. This will be apparent from the aforementionedaddress assignment of the block shown in FIG. 4.

In the simultaneous test mode of 64 bits, as will be describedhereinafter, the addresses Y2 and Y3 are also restricted. If, however,the fuses corresponding to the addresses Y2 and Y3 are eliminated,sixteen redundancy YS lines are simultaneously formed in one block.Specifically, redundancy data lines of 16×4 (the number of I/O) 64 pairsare simultaneously remedied so that the efficiency is deterioratedbecause such a large number of redundancy data lines have to beprepared. For the addresses Y2 and Y3, therefore, only the YS linescorresponding to the complementary data lines having an actual defectwhen in the simultaneous test of 64 bits are changed to the redundancyYS lines, whereas the remainder selects the normal YS lines (i.e., themulti-selection of the 4 YS lines due to the restriction of theaddresses Y2 and Y3). As a result, there is no necessity for preparingfour times of the redundancy data lines despite of the test mode of 64bits of the YS multi-selection type.

Since the YS lines extend over the plural blocks, as described above,the data lines are troubled in the plural blocks sharing the YS lines ifa defect is caused in the YS lines. If the redundancy decoder isassigned to each block so as to remedy that trouble, the number ofredundancy decoders is increased to drop the remedy efficiency. In orderto prevent this, two fuses are attached to each of the block dividingaddresses X8 to X11, and the lower fuse FUSE is cut. Then, thecorresponding X-addresses are not compared. If the lower fuses FUSE ofthe addresses X8, X9 and X11, for example, are cut, the eight blockssharing one YS line are restricted and can be remedied by one redundancydecoder to improve the efficiency. For a defective sense amplifier,likewise, the righthand and lefthand data lines of the sense amplifiercan be remedied by one redundancy decoder if only the lower fuse FUSE ofthe address XB is cut.

In FIG. 46, the upper circuit corresponds to the enabling circuit, andthe lower circuit corresponds to the addresses Y4 to Y9. In FIG. 47, theupper circuit corresponds to the addresses Y2 and Y3, and the lowercircuit corresponds to the addresses X8 to X11.

The fuse FUSE is initialized through a MOSFET which is turned ON by thesignal FUS of one-shot pulse. If the fuse FUSE is cut, it is fixed atthe ground potential by a MOSFET which is turned ON by the high outputlevel of the inverter circuit If the fuse FUSE is not cut, the input ofthe inverter circuit is fixed at the high level.

During the remedy, a signal RDJ takes the high level, if the addressprogrammed by the redundancy decoder is coincident to the input address,but takes the low level in the case of inconsistency. In the operationother than the remedy, the signal RDJ is fixed at the low level.

In the simultaneous test of 64 bits, a signal YMB takes the low level,and signals YFIJ and YFIJB output the states of the fuse correspondingto the addresses Y2 and Y3. These addresses Y2 and Y3 are not compared(but restricted). In the test of the redundancy data lines, theaddresses X8 to X11 are restricted. When the addresses Y2 and Y3 are inthe states of (0, 0), (1, 0), (0, 1) and (1, 1), the redundancy decodersof J=0, 3, 6 and 9 are remedied to correspond to the four redundancy YSlines. This structure is similar to that of the aforementionedX-redundancy circuit.

In FIG. 48, there are formed redundancy YS line selection signals YRD0Bto YRD3B which correspond to signals RD0 to RD2, RD3 to RD5, RD6 to RD8,and RD9 to RD11, respectively.

The signal YRD inhibits the selection or the normal YS lines when in theredundancy selection, if it is at the high level. For the simultaneoustests of 64 bits, however, the signal YRD is fixed at the low level inresponse to the low level of a signal YMB so that the normal YS linesare simultaneously selected Signals RA0JB to RA3JB monitor the states ofthe fuses FUSE corresponding to the addresses Y2 and Y3. In the normalmode, the signals RA0JB to RA3JB are fixed at the high level in responseto the high level of the signal YMB. In the simultaneous test of 64bits, the states of the fuses of the addresses Y2 and Y3 are decoded inresponse to the high level of the signal RDJ, when the remedy address isselected, to drop one of the outputs to the low level (in a manner tocorrespond to the pre-decode signals of the defective addresses Y2 andY3).

The signals RY20B to RY23B adopt the OR logic of J=0 to 11 so that twoor more of the signals RY20B to RY23B can be dropped to the low levelwhen the addresses of the twelve redundancy decoders of J=0 to 11 exceptthe address Y2 and Y3 happen to be coincident so that they are remedied.In other words, in case two of the four YS lines to be restricted by theaddresses Y2 and Y3 are remedied, they are distributed to the redundancyYS lines, whereas the remaining two are distributed to the normal YSlines.

In order to check the redundancy YS lines, namely, to select theredundancy YS lines in the test mode thereby to accomplish thewrite/read tests in the memory cells of the redundancy YS lines, theredundancy YS lines (e.g., YSR0 to YSR3) have to be selected for thedesignation of any address of the address signals X8 to X11. Moreover,the two bits of the address signals Y2 and Y3 are used for designatingthe redundancy YS lines. Specifically, the STB (i.e., the redundancytest signal) signal or the VCC signal is fed in response to signals BI(I=2 and 3) and A (corresponding to the redundancy decoders of L=8, 9,10 and 11). As a result, the fuses of the defective address are cut notactually but equivalently by the address signals in the aforementionedtest mode so that the redundancy YS lines of the designated addressescan be selected. Since this circuit is basically similar to theaforementioned redundancy circuit of the X-system, the detaileddescriptions of the individual signals will be omitted.

The defect remedying method according to the present invention will bedescribed from another view point in the following.

FIG. 91(A) is a conceptional diagram for explaining one example of thedefect remedy in the simultaneous test mode of multiple bits by theaforementioned multiplex selection of the Y-system.

In the same Figure, the abscissa designates the X-address, and thecoordinate designates the Y-address. In case a RAM is constructed tohave a storage capacity of about 16 Mbits like this embodiment, theX-axis has 4,096 addresses, and the Y-axis also has 4,096 addresses. Inthe defect remedying technology of the prior art, a switching is made tothe redundancy circuit in response to one defective address in the X-and Y-directions. If, therefore, one of the Y-addresses is defective, anaccess to its bit lines to be coupled to the 4,096 memory cells isinhibited, and a switching to the redundancy bit lines to be coupled tothe 4,096 memory cells is accomplished. Since the scale of theredundancy circuit is thus enlarged, the memory cells in the shownembodiment are grouped into sixteen memory blocks by dividing the X- andY-addresses into four blocks while using the higher two bits of the X-and Y-addresses so that the data lines can be designated at the unit ofeach block.

On the other hand, the Y-system is multiplexly selected in theaforementioned simultaneous test of multiple bits or when the higher 2bits of the Y-addresses are restricted to make a structure of ×4 bits.In case there is any defect, therefore, the defect remedying method ofthe prior art changes all to the redundancy circuits. Then, it isnecessary to change the bit lines having no defect to the redundancy bitlines only for the multi-selection test or for the structure of ×4 bits.When the Y-system is to be selected simultaneously for the fouraddresses, as shown in the same Figure, only the block having the defectbit lines and the YS selection lines is switched to redundancy bit linesRBL, and the bit lines corresponding to the remaining three addresses tobe selected simultaneously are the normal bit lines NBL. Incidentallywith this block structure, the other memory blocks to be divided by theX-addresses have their bits lines left unselected. Thanks to thisstructure, only the defective ones are switched to the redundancy bitlines so that the redundancy bit lines to be prepared can be drasticallyreduced.

FIG. 91(B) is a conceptional diagram for explaining another embodimentof the defect remedy of bit lines when in the aforementioned normalmode.

In the example of the same Figure (B), only those of the bit linesbelonging to the same Y-address, which are shared by a defective one ofthe four blocks divided by the X-addresses, are changed to theredundancy bit lines RBL, whereas the normal bit lines NBL are selectedfor the other blocks. By the defect remedy of such block unit, it ispossible to reduce the number of redundancy bit lines or YS selectionlines to be prepared.

FIG. 91(C) is a conceptional diagram for explaining another embodimentof the defect remedy of word lines in the aforementioned normal mode.

In the example of the same Figure (C), only those of the word linesbelonging to the same X-address, which are shared by a defective one ofthe four blocks divided by the Y-addresses, are changed to theredundancy word lines RWL, whereas the normal word lines NWL areselected for the other blocks. By the defect remedy of such block unit,it is possible to reduce the number of redundancy word lines to beprepared. However, the DRAM, to which the X-address signals of thisembodiment are multiplexly inputted prior to the Y-address signals,cannot use these Y-address signals as they are. Therefore, the defectremedying method like the aforementioned one can be realized byprogramming the addresses called the block addresses equivalent to theY-addresses by the aforementioned fuse means.

FIG. 49 presents circuit diagrams showing portions of one embodiment ofthe predecoder circuits of the Y-system including a circuit for formingthe selection signals of the main amplifiers.

A signal ASK (AS0 to AS3) selects a group of main amplifiers (i.e., oneof four pairs of I/O lines). Signals AY20U/D to AY23U/D predecode theaddresses Y2 and Y3. The upper and lower mats are divided by the addressX10. In the simultaneous test of 64 bits, the. predecodes of theaddresses Y2 and Y3 are ignored so that the signals RY20B to RY23B ofFIG. 48 are outputted as they are.

Signals Y0UB to Y3UB and Y0DB to Y3DB are predecode signals, which areoutputted in response to the signal YP by predecoding the addresses Y 4and Y5, and are set to have the data line selection timing. The resettiming is regulated by the signal CE. The signals Y0UB to Y3UB and Y0DBto Y3DB take the high level, when the signal YRD is at the high level,to inhibit the selection of the normal YS lines.

In the simultaneous test of 64 bits, the four signals AY20U/D to AY23U/Dtake the high level so that the four YS lines are selected, if the fourYS lines to be restricted by the addresses Y2 and Y3 are not remedied.If the YS lines are remedied, on the contrary, one to four of thecorresponding signals AY20U/D to AY23U/D are not outputted, but one tofour redundancy YS lines are selected so that the redundancy YS linesand the normal YS lines are simultaneously selected. The signals AY60U/Dto AY83U/D are the pre-decode signals of the addresses Y6 to Y9. SignalsYR0U/DB to YR3U/DB select the redundancy YS lines and correspond tosignals Y0U/DB to Y3U/DB.

FIG. 50 presents circuit diagrams showing the unit circuits of theY-decoders and the redundancy YS line selection circuit.

The aforementioned pre-decode signals are decoded by a tri-input NANDgate circuit. The decoded outputs and the Y-selection timing signal YKUB(K=0 to 3) are fed to the NOR gate circuits to form column selectionsignals YS0 to YS3 from the respective NOR gate circuits. The redundancycolumn selection signals YSR0 to YSR3 are formed from the signals whichare formed by the aforementioned redundancy decoder circuits.

FIG. 51 presents circuit diagrams showing one embodiment of the nibblecounter.

In the normal mode, an address signal NAK corresponding to the internaladdress signal BYI is outputted. In the nibble mode, the internaladdress signal BYI of the first cycle is counted up at first. When thememory access is to be accomplished by the structure of ×4 bits, thesignal NAK is fixed at the high level (VCC) by the master slice in theswitch mode.

FIG. 52 presents circuit diagrams showing one embodiment of the controlcircuit for generating the control signals of the Y-system.

A signal MA is one for controlling the main amplifier operations. Asignal DS is one for controlling the output of the data of the mainamplifiers. The signal MA is generated in accordance with the generationof a signal AC1B (RYP). A signal R1 determines the reset timing of themain amplifiers.

The signal DS is generated in response to the signal MA and is reset bythe signals C1 and R1. Specifically, the control of the data output ofthe main amplifiers is reset when both the signals RAS and CAS are atthe high level.

A signal WR is one for discriminating the read/write operations. Theinitial step is controlled by the signal R1 to reduce the currentconsumption in the standby state.

A signal DOE is one for controlling the data output buffer and isgenerated when in the read mode. In the case of the structure of ×1 bit,the signal DOE is generated by the logical product of the signals C1 andWR. In the case of the structure of ×4 bits, the signal DOE is generatedby the logical product of the output enable signals OE, C1 and WR. Acontrol signal OE is latched by the signal DL of the WE system for atime t_(OEH) (i.e., the holding time period of the signal OE from thesignal WE).

FIG. 53 presents circuit diagrams showing one embodiment of theoperation mode deciding circuit.

Signals RN and RF and signals WN and WF control the normal operations,the CBR operations and the WCBR operations. The signals RN and RFcontrol the signals CE adn YE, and signals CRB and LFB control thetesting circuits, i.e., the set/reset of the addresses of the WCBRoperations.

FIG. 54 presents circuit diagrams showing portions of the embodiment ofthe control circuit of the Y-system.

The signal YL causes the Y-address buffer shown in FIG. 45 to latch theaddresses. The generation timings are different for the individualoperations modes, as has been described hereinbefore. One example of theoperation waveforms is shown in FIG. 77.

In response to the fast page mode (i.e., normal mode), the Y-addressesare latched in synchronism with the CAS. In the nibble mode, theY-addresses are latched for the RAS cycle. This is because the addresssignals are generated by the nibble counter in the nibble mode. In thestatic column mode, the Y-addresses are latched when in the writeoperation. In the counter test mode of the CBR, the Y-addresses arelatched. In the WCBR mode, the Y-addresses are latched for the RAS cycleperiod.

The signal DL controls the set-up/hold of the data of the data inputbuffer. In the fast page mode or nibble mode, the setting isaccomplished when the CAS and the WE are at the low level, and theresetting is accomplished when the CAS is at the high level. In thestatic column mode, the setting is accomplished when the CAS or WE is atthe low level, and the resetting is accomplished at the end of the writeoperation.

A signal OLB is one for latching the written data so that they may notbe outputted to the DO. This signal corresponds to the read-modify-writeoperations, which in turn correspond to the time period t_(WOH) (i.e.,the output hold time period from the signal WE) in the static columnmode.

FIGS. 55 and 56 present diagrams showing the circuits of the portions ofthe control circuit of the WE system.

In FIG. 55, the WE (Write Enable) signal is fed to the input circuitconstructed of the CMOS inverter circuit. This CMOS inverter circuit forthe input buffer is made to have the logic threshold voltage such asabout 1.6 V like before. The power source voltage VCC for the peripheralcircuit of the DRAM of this embodiment is set at 3.3 V twice as high asthe aforementioned logic threshold voltage of 1.6 V and corresponds tothe signal at the TTL level.

Signals W1 and W2 are used for controlling the write operations. In thestandby state, the signals W1 and W2 are dropped to the low level. Inthe operation, the signals W1 and W2 are changed in synchronism with thechanges in the signal WE. The signal W1 accomplishes the RAS/WE logiccontrol (WN/WF), and the signal W2 accomplishes the CAS/WE logiccontrol. The write setting is delayed to retain the time period t_(ASC)(i.e., the column address set-up time period). A signal W3B is aone-shot pulse formed by the signal W2 to form a signal W4B.

In FIG. 56, a signal WYP performs the control till the write signal istransmitted from the data input buffer to the input/output lines I/O,and a signal WYPB performs the control till the write signal istransmitted from the input/output lines I/O to the bit lines.

A signal IOU precharges the input/output lines after the writeoperations so as to cope with the subsequent read cycle. A signal WLlatches the addresses and data when in the static column mode. FIG. 76is a timing chart showing one example of the write operations.

FIG. 57 presents circuit diagrams showing one embodiment of the datainput buffer.

An input circuit is constructed of a NAND gate circuit and has a logicthreshold voltage similar to that of the aforementioned other inputcircuits. In the structure of ×1 bit, a control signal A of this gate issubstantially invalidated when one of the four input buffers takes thesignal R1 whereas the remaining three input buffers are fed with theground potential VSS of the circuit. In the structure of ×4 bits, on theother hand, the signal A is the signal R1 in response to all the fourinput buffers. The reason why the NAND gate circuit is used at the inputbuffers to be operated and is fed with the signal R1 is to reduce thecurrent consumption in the standby state like before. The set-up/hold ofthe write data are controlled by the signal DL.

A signal MKI is used to control the write mask mode in the structure of×4 bits. The write/non-write are controlled by the data of the signalsDQ1 to DQ4 when the signal RAS is set. A signal DI (0 to 3) is furtherdivided at the unit of nibble address NAI.

FIG. 58 presents circuit diagrams showing one embodiment of the controlcircuit of the main amplifier, and FIG. 59 is a circuit diagram showingone embodiment of the main amplifier.

A signal RMA is a timing signal for controlling the operation of themain amplifier. A signal WMA control the signal transmission (or thewrite operation) from the data input buffer to the input/output linesI/). Signals ILAij to ILCij pull up the input/output lines I/O, and asignal IOU is one for shorting the input/output lines I/O.

In the normal mode, one main amplifier is operated by the signal RMA. Inone test mode, sixteen main amplifiers are brought altogether intooperative states to compare all of 16 bits in response to a signal TE.In another,test mode, moreover, 64 bits are compared altogether by themultiple selection of the YS lines in response to the signals TE andYMB. FIG. 89(A) is a circuit diagram for explaining the principle of themulti-bit test by taking the 4-bit parallel test using a pair of mainamplifiers as an example. In accordance with the example of the sameFigure, the aforementioned sixteen main amplifiers are grouped intoeight pairs. The read data of total 8 bits are sent out through the twoI/O pairs of 4 bits, which are multi-selected by the four YS lines ofevery two I/O pairs corresponding to the pair of the main amplifiers, inparallel to the aforementioned eight pairs of main amplifiers so thatthe multi-tests of total 64 bits are accomplished.

Further descriptions will be made with reference to FIG. 89(A). One ofthe paired main amplifiers MA has its input connected commonly withcomplementary bit lines BL1 and BLB1 to BL4 and BLB4, which correspondto read signals of 4 bits, through Y-switch MOSFETs and input/outputlines I/O and I/OB, respectively. The other of the aforementioned pairedmain amplifiers MA has its input fed with a reference voltage VR. Thisreference voltage VR is set at an intermediate level between the readsignal at the high level and the signal at incoincidence of 1 bit, asshown in the waveform chart of FIG. 89(B). If the complementary bitlines BL1 and BLB1 are at the logic “0” (namely, the BL1 is at the lowlevel “L” whereas the BLB1 is at the high level “H”), as shown, thelevel of the input/output lines I/O is dropped to a lower level, asindicated by the broken line of the same Figure, to the extent which iscaused by Y-switch MOSFET (M2) and the sense amplifier MOSFET (M3) areconnected to the pull-up MOSFET (M1). Therefore, the aforementionedreference voltage VR is set at the intermediate level between the highlevel and the low level for the incoincidence of 1 bit by connecting thetwo Y-switch MOSFETs (M2) and the two sense amplifier MOSFETs (M3) inseries with the aforementioned pull-up MOSFET (M1). In the embodimentshown in FIG. 89, therefore, the whole bit logic “1” is written. If thelogic “0” of 1 bit is present, the output signals of the main amplifierscorresponding to the input/output lines I/O are changed from the highlevel to the same low level as that of the outputs of the mainamplifiers corresponding to the input/output lines I/OB so that theerrors are detected. If, on the contrary, the logic “0” is written inand read out from all the 4 bits, the input/output lines I/OB take thehigh level when the whole bit logic “0” is read out. If there is aninconsistency of 1 bit, the level of the input/output lines I/OB isdropped down like before so that the output signals of the mainamplifiers corresponding to the input/output lines I/OB of the pairedmain amplifiers are changed from the high to low levels equal to that ofthe outputs of the, main amplifiers corresponding to the input/outputlines I/O, so that the errors are detected. Incidentally, if all thebits are coincident, the outputs of the paired main amplifiers aredivided into the high and low levels.

In this multi-bit test, as in the state of FIG. 89, the input/outputlines I/OB have a tendency of taking a relatively low level because theyare fed with the low level of the outputs of the three sense amplifiers.As a result, there arises a fear that the low level of the input/outputI/OB is transmitted to the bit line BLBl which has been subjected to adefective reading so that it may possibly be written with the normaldata by inverting the sense amplifiers.

As a counter measure for this, the conductance of the pull-up MOSFET(M1) is increased when in the aforementioned multi-bit test mode. Inthis mode, more specifically, there is provided the pull-up MOSFET whichis to be turned ON in response to that signal. Thus, the drops of thelow level of the input/output lines I/O and I/OB can be reduced toprevent the aforementioned erroneous writing.

In the aforementioned multi-bit test, on the other hand, the operationvoltage is switched from the VCC to the VCCE of about 5 V or the boostedvoltage VCH by the switch MOSFET which is to be turned ON in response tothe control voltage. According to this structure, the level of theinput/output lines can be raised to an extent corresponding to theaforementioned voltage switching so that the aforementioned erroneouswriting due to the low level can be prevented.

On the other hand, the threshold voltage of the pull-up MOSFET can bedropped to the low threshold voltage so that the pull-up (or bias) levelof the input/output lines may be accordingly raised. In the operation atthe low voltage VCC such as about 3.3 V like this embodiment, thelow-level margin for preventing the erroneous writing can be reducedbecause of the low pull-up level, if the threshold voltage of thepull-up MOSFET is high.

In the embodiment shown in FIG. 54, the two True and Bar I/O line pairsto be intrinsically connected with the aforementioned two mainamplifiers are connected to each other so that the two main amplifiersare commonly used in the aforementioned mode. This prevents the numberof main amplifiers from being doubled. The aforementioned eight pairs ofmain amplifiers compare the total 8 bits, i.e., 4 bits for each of theI/O line pairs so that 64 bits are simultaneously tested.

By adopting the aforementioned multi-bit test, it is possible to shortenthe test time period of the RAM having a high storage capacity such asabout 16 Mbits.

In the write mode, the signals from the data input buffers are fed tothe aforementioned input/output Lines I/O in response to the signal WMA,and the data are written in the main amplifiers in response to thesignal RMA. This corresponds to the nibble mode and the fast page mode.

FIG. 60 is a circuit diagram showing one embodiment of the outputcontrol circuit of the data of the main amplifiers.

One pair of main amplifier output groups MAi0 to MAi3 and MAi0B to MAi3Bis selected by the main amplifier selection addresses AS0 to AS3, andthe output groups selected by a nibble address NAi are sent out tooutput lines MOiB and MOi in response to the signal DS. Thus, one ofsixteen main amplifiers is selected. At the output of the unit of ×4bits, the nibble address NAi is fixed at the high level.

The signal DS is reset in response to the RAS and CAS resets when in thefast page mode. In the nibble mode, the signal DS is left at the highlevel because it is sufficient to feed the data to the four mainamplifiers in the first cycle and to output the data from the mainamplifiers in the second cycle.

In the test mode for forming the signal TE, the data of the four mainamplifiers are gathered to one output signal MOi through the comparator(or NAND gate).

FIG. 61 presents circuit diagrams showing one embodiment of the outputcontrol circuit of the main amplifiers.

The signal OLB controls the data output to the data output buffer. Thedata latch is accomplished in the read-modify-write. In response to thesignal TE, all the sixteen main amplifiers are activated in the testmode to output the data to output signals MO0 to MO3 and MO0B to MO3B.These comparison output system may be binary or ternary.

In the binary system, the high- and low-levels are outputted to theoutput DO/DOB for all logic “1” or logic “0”, and the low- andhigh-levels are outputted for failure. In the ternary system, the high-and low-levels are outputted to the output DO/DOB for all logic “1”, andthe low- and high-levels are outputted for all logic “0”. In the case offailure, the low- and low-levels are outputted.

The aforementioned binary output system is established if a signal TW isat the high level, and the aforementioned ternary output system isestablished if the signal TW is at the low level.

FIG. 62 is a circuit diagram showing one embodiment of the data outputbuffer.

The data output buffer is equipped at its input portion with a levelconverter circuit. As has been described hereinbefore, the internalcircuit is operated by the dropped voltage VCC. Therefore, the read datatransmitted through the main amplifiers are generated in response to theoperation voltage VCC. The data having passed through the NAND gatecircuit in response to the signal DOE have their level converted by theNOR gate circuit of the latch mode, which is operated by the powervoltage VCCE fed from the outside. Since the push-pull output unitcomposed of the N-channel MOSFET is driven by such level convertercircuit, the output level at the high-level side can be raised, and theamplitude of the drive signal can be enlarged to speed up theoperations.

The aforementioned output portion is equipped with MOSFETs andresistance elements for controlling the gates of the output MOSFETs. Thethreshold voltage of the MOSFET, which is connected between the gate andsource of the output MOSFET at the side of the power voltage VCCE andhas its gate supplied steadily with the ground potential VSS, is madelower than that of the output MOSFET. As a result, when an outputterminal DOUT takes a negative potential, the MOSFET having the lowerthreshold voltage is turned ON to short the gate and source of theoutput MOSFET. As a result, the output MOSFET is not turned ON by theaforementioned negative voltage.

There is also provided an output circuit which is operated with therelatively fast timing having passed through the aforementioned outputgate circuit, to quicken the rising and breaking timings of the outputsignal. Thus, the level is changed to the value which is regulated bythe output circuit made receptive of the data having passed through thelevel converter circuit. Thanks to this structure, the output level canbe linearly changed for a relatively long period while speeding up theoperations, so that the level of the noises to be generated in the powerlines or ground lines by the level change of the output signals can bereduced.

FIGS. 63 and 64 are circuit diagrams showing one embodiment of the testcircuit.

With the timing WCBR, the test function is set. By this WCBR, there isoutputted the test signal which corresponds to the address taken in. Thesignal LFB is formed by the WCBR so that the external address signalscan be taken in.

A signal FR resets all at the logic “0” when the power supply is made.

The reset of the test function is accomplished by resetting all theaddresses at the logic “0” such that the signal FR is raised to the highlevel for the precharge period of the RAS signal for the RAS onlyrefresh cycle and the CBR refresh cycle.

In the test mode, the following modes are prepared for signals FMNBwhich are formed by combining the AFI to AFL of 4 bits corresponding tothe address signals Y0 to Y3. The modes are for: (1) the tests of ×16bits; (2) the tests of ×64 bits; (3) the switching from the internalvoltage VCC to the external voltage VCCE; (4) the monitor of theinternal voltage VCC; (5) the monitor of the internal voltage VDL; (6)the refresh of 2,048 (or the operation of the 8,192 bits); (7) theredundancy area test; and (8) the speed-up test.

FIG. 65 presents circuit diagrams showing one embodiment of the controlcircuit for designating the operation modes.

By selecting the high-/low-levels and the high impedance for the bondingpads FP0 and FP1, the following modes are set from the selectedcombination in accordance with the structures of ×1 bit and ×4 bitsdesignated by the aluminum mat slice.

In the structure of ×1 bit, both signals SC and NB are dropped to thelow level to designate the fast page mode when both the pads FP0 and FP1are at the high impedance. If the pad FP0 is at the low level whereasthe pad FP1 is at the high impedance, the signal SC takes the high levelto designate the static column mode. If the pad FP0 is at the highimpedance whereas the pad FP1 is at the high level (VCCE), the signal NBtakes the high level to designate the nibble mode.

In the structure of ×4 bits, when both the pads FP0 and FP1 are at thehigh impedance, both the signals SC and NB take the low level todesignate the fast page mode. If the pad FP0 is at the low level whereasthe pad FP1 is at the high impedance, the signal SC takes the high levelto designate the static column mode. If the pad FP0 is at the highimpedance whereas the pad FP1 is at the high level (VCCE), the signal WBis formed to cause the write mask mode with the fast page mode. If thepad FP0 is at the low level whereas the pad FP1 is at the high level(VCCE), the signal WB is likewise formed to cause the write mask modewith the static column. In the write mask mode, the pin to be writtenfrom the output terminal I/O can be set by setting the WE signal at thelow level when the RAS signal breaks.

FIG. 66 presents circuit diagrams showing one embodiment of anothercontrol circuit.

The signal WKB monitors of the level of the bias voltage VBB of thesubstrate. When this substrate bias voltage VBB drops to −0.7 V or less,the signal WKB takes the low level. If the substrate bias voltage VBB isshallow, the threshold voltage of the MOSFET drops so that a relativelylarge through current is caused to flow by the circuit operationsthereby to make the latch-up liable to occur. Therefore, the access tothe RAM is inhibited by the high level of the signal WKB.

A signal INT monitors the level of the power voltage VCCE. The signalINT is dropped to the low level for the voltage VCCE>3V. In other words,the internal initial state is set in response to the signal INT when theexternal power voltage is low.

In this embodiment, the structure of the delay circuit, as indicated bya block box, is specifically shown. If a terminal SET is raised to thehigh level (VCC), the amount of delay can be shortened. This concept canbe widely used for adjusting the timing of the RAS system and forgenerating the pulses of the CAS and WE system.

An output terminal Q/DQ4 is used as a monitor terminal of the internalvoltage. With the data output buffer coupled to the monitor terminalbeing in the high impedance state, the operation voltage VCC for theperipheral circuit is outputted through the MOSFET to be switched by asignal VMCH, and an operation voltage VDL for the sense amplifier isoutputted through the MOSFET which is switched by a signal VMDH.

On the other hand, the output terminal Q/DQ4 is also used as a signatureterminal for deciding the presence of the defect remedy. In the chiphaving its defect remedied, the SIGB takes the low level. The chip ofdefect remedy is decided in view of the fact that the current flows intothe ground potential of the circuit in case a voltage three times ormore of the threshold voltage Vth as high as the VCCE is applied to theQ/DQ4 terminal.

FIG. 67 presents circuit diagrams showing one embodiment of thesubstrate back bias voltage generating circuit.

In this embodiment, the low voltage VCC for the peripheral circuit isused as the operation voltage. The substrate back bias voltage is thusformed by the internal voltage VCC because the internal voltage VCC isstabilized, as will be described in the following, so that the substratebias voltage can be stabilized.

The substrate bias voltage VBB is formed by bias voltage generators VBBAand VBBS. The substrate bias voltage generator VBBA is a main generatorwhich operates for compensating a substrate current I_(BB) when thesubstrate level is shallow and due to the circuit when in the operation.The substrate bias voltage generator VBBS is a subsidiary generatorwhich steadily operates for compensating the fluctuations of the VBB dueto the leak current and the fine DC current.

A signal VBSB is a monitor output at the level of the substrate voltageVBB. The aforementioned oscillator is controlled by the signal VBSB tooperate till the VBB is dropped to about −2 V by the circuit VBBA whenthe aforementioned substrate level is shallow.

A terminal VBT is used to interrupt the operations of the circuits VBBAand VBBS so that it may set the substrate voltage through the VBB padfrom the outside to evaluate the operation margin.

FIG. 68 presents circuit diagrams showing one embodiment of the internalboost voltage generator.

A circuit VCHA is a main boost voltage. When the level is dropped by themonitor signal VHSB of the boost voltage VCH or when the RAM is accessedto by the signal RIB, the aforementioned boost voltage VCH such as about5.3 V is generated by the charge pump circuit which is made receptive ofthe internal operation voltage VCC for the peripheral circuit and anoscillatory signal OSCH formed by the oscillator. A circuit VCHS is asubsidiary boost voltage generator which steadily operates to form theboost voltage VCH. This circuit VCHS has such a small current supplyability as can compensate the leakage current of the word lines.

For the acceleration tests, as will be described later, the internalvoltage VCC is raised accordingly as the power voltage VCCE is raised toa constant level or more. In response to this, the boost voltage VCH isalso raised to a constant level in accordance with the rise of thevoltage VCC. This level is clamped by the MOSFET of the diode mode whichis disposed at the output portion.

The terminal VHT is provided to interrupt the operations of the circuitsVCHA and VCHS so that it may set the boost voltage through the VCH padfrom the output side to evaluate the operation margin. Although notshown, the capacitor for dropping the power impedance of the boostvoltage VCH is dispersed for the unit of the operating circuit such aseach memory mat.

FIG. 69 presents circuit diagrams showing one embodiment of the internalvoltage drop circuit.

A reference voltage VREF is a highly precise reference voltage which isformed by making use of the difference between the threshold voltagesVth of the MOSFETs. From this voltage, the constant voltage VL is formedand amplified in series by the operation amplifier to generator theaforementioned voltages VDL and VCC of about 3.3 V. In order to reducethe operation current, the circuits for generating the voltages VCC andVDL, respectively, are operated only when the DRAM is brought into itsoperating state by signals LD and LC. There is provided another circuitwhich is steadily brought, when the power voltage VCCE is at a constantlevel or more, into its operating state to form the drop voltage for thestandby by a signal LS.

Immediately after the power is made, a signal SB is formed by the signalINT till the external voltage VCCE reaches a constant level. Inaccordance with this, the signals LD, LC and LS are formed to bring allthe circuits into operative states to raise the internal circuitoperating voltage at a high speed.

In the same Figure, the circuits composed of resistors and capacitorsare used for enlarging the phase margin for preventing the oscillations.

Fuses F1 to F4 can be selectively cut with a laser beam to adjust thereference voltage VL.

In the test function, the signals LD, LC and LS are dropped to the lowlevel by a signal VE to interrupt the operations of the operationamplifier, and the low level is fed to the gate of the P-channel outputMOSFET of the arithmetic amplifier to turn ON the MOSFET by the MOSFETwhich is to be turned ON by a signal VHE. As a result, the internalvoltages VDL and VCC are switched by and to the external voltage VCCEthrough the P-channel MOSFET which is turned ON.

If, moreover, the external power voltage VCCE exceeds a constant level(e.g., about 6.6 V), the reference voltage VL accordingly rises to raisethe internal voltages VCC and VDL. This corresponds to the accelerationtest such as the aging.

FIG. 70 is a timing chart showing one example of the operations of theRAS system.

In the same Figure, there are shown schematic waveforms of the majortiming signals for selecting and resetting the word lines WL from thestart of the memory accesses by the RAS signals.

FIG. 71 is a timing chart showing one example of the operations of theRAS system.

In the same Figure, there is shown the timing chart for selecting theword lines. In the second cycle, there is also shown the redundancytiming.

FIG. 72 is a timing chart showing one example of the operations of theRAS system.

In the same Figure, there are shown timing signals for activating thesense amplifiers and waveform charts to be driven by the former.

FIG. 73 is a timing chart showing one example of the operations of theX-address buffers.

In the same Figure, there are shown the mutual timings between the RASsignals and the CAS signals.

FIG. 74 is a timing chart showing one example of the operations of theCAS system.

In the same Figure, there are presented waveform charts of the majorsignals in the order of the read mode (READ), the early write mode (EW),the read-modify-write mode (RMW), the RAS only refresh mode, the CBRrefresh mode, the counter test mode and the test mode set (WCBR).

FIG. 75 is a timing chart showing one example of the address sectionoperations of the CAS system. In the same Figure, there are shown themajor timing-signals for the address selections of the Y-system.

FIG. 76 is a timing chart showing one example of the write operations.

In the same Figure, there are shown the major timing signals of the WEsystem.

FIG. 77 is a timing chart showing one example of the operations of theY-address buffers.

In the same Figure, there is centered the timing signal YL forcontrolling the address latches in the fast page mode (FP), the nibblemode (N), and the static column mode (SC).

FIG. 78 is a timing chart showing one embodiment of the operations ofthe test mode.

In the same Figure, there are centered the address taking and latchingoperations.

FIG. 79 is a timing chart showing one example of the operations of theCAS system.

In the same Figure, there are exemplified waveform charts of theindividual signals of the test mode system in the order of the read mode(READ), the early write mode (EW), the read-modify-write mode (RMW), theRAS only refresh mode, the CBR refresh mode, the counter test mode andthe test mode set (WCBR).

FIG. 80 is a timing chart showing one example of the operations of theCAS system.

In the same Figure, there are exemplified waveform charts of theindividual signals of the structure of ×4 bits in the order of the readmode (READ), the early write mode (EW), the read-modify-write mode(RMW), the RAS only refresh mode, the CBR refresh mode, the counter testmode and the test mode set (WCBR).

FIG. 81 is a timing chart showing one example of the operations of theCAS system.

In the same Figure, there are exemplified waveform charts of theindividual signals of the write mask mode in the order of the read mode(READ), the early write mode (EW), the read-modify-write mode (RMW), theRAS only refresh mode, the CBR refresh mode, the counter test mode andthe test mode set (WCBR).

FIG. 82 is a block diagram showing another embodiment of the defectremedying method according to the present invention.

For a plurality of word lines to be selected by an X-decoder (includingthe word line driver), there is provided one redundancy word line. Thisredundancy line is arranged to intersect the plural word lines at theportion corresponding to the X-decoder, i.e, in parallel with the rowsof the output terminals of the X-decoder. The redundancy word lineintersects the plural word lines to be remedied by the two parallelwiring lines, although not especially limitative. These two parallelwiring lines have their one-side ends supplied with the groundpotential.

In this structure, the word lines are steadily in non-selected statebecause the redundancy word line is supplied with the ground potentialwhen the word lines have no defect.

If one word line is defective (e.g., cut) at a portion indicated by aletter X, the word line is cut at a portion indicated by a symbol oftriangle. Likewise, the redundancy word line is so cut at the righthandside (at the side of the redundancy word line) from the defective wordline as is indicated by the triangle so that it may be isolated from theground potential. Moreover, the decode output forming the selectionsignal of the defective word line is connected with the redundancy wordline at the portion of intersection, as indicated by a symbol of circle.Likewise, in order to bring the defective word line into thenon-selected state. The aforementioned cutting and connecting operationsof the wiring lines are accomplished by making use of the wiring workingtechnology using the laser beam, although not especially limitative.

In this structure, the defective word line is cut from the outputterminal of the word line selection circuit and is replaced by theredundancy word line, thus requiring neither the memory circuit forstoring the defective address nor the address comparator. This makes itpossible to highly integrate and reduce the power consumption of thesemiconductor memory device. Since none of the aforementioned addresscomparison is required, the memory access can be speeded up.

In case, on the other hand, the aforementioned redundancy word line isprovided for a plurality of word lines, it is steadily supplied, if notused, with the ground potential to establish the shield action forsuppressing the coupling of the word lines.

FIG. 83 is a block diagram showing another embodiment of the defectremedying method according to the present invention.

One redundancy column selection line is provided for a plurality ofcolumn selection lines which are formed by the Y-decoder circuit. Thesecolumn selection lines are connected with the gates of the column switchMOSFETs, which are contained in the sense amplifiers, to connect theshown bit (or data) lines selectively with the common input/outputlines. The redundancy column selection line is arranged to intersect theaforementioned plural column selection lines at the portionscorresponding to the Y-decoder, namely, to extend in parallel with therows of the output terminals of the Y-decoder. The aforementionedredundancy column selection lines intersect the plural column selectionslines to be remedied by the two parallel wiring lines. These twoparallel wiring lines have their one-side ends supplied with the groundpotential.

In this structure, the bit lines and the sense amplifiers are steadilyin non-selected state because the redundancy column selection issupplied with the ground potential when they have no defect.

If one bit line is defective (e.g., cut) at a portion indicated by aletter X, the column selection line is cut at a portion indicated by asymbol of triangle. Likewise, the redundancy column selection line is socut at upper side (at the side of the redundancy column selection line)from the column selection line corresponding to the defective bit lineas is indicated by the triangle so that it may be isolated from theground potential. Moreover, the decode output forming the selectionsignal of the defective bit line is connected with the redundancy columnselection line at the portion of intersection, as indicated by a symbolof circle. Likewise, in order to bring the column selection linecorresponding to the defective bit line into the non-selected state. Theaforementioned cutting and connecting operations of the wiring lines areaccomplished by making use of the wiring working technology using thelaser beam, although not especially limitative.

In this structure, the column selection line corresponding to thedefective bit line is cut from the output terminal of the Y-decoder andis replaced by the column selection line corresponding to the redundancybit line, thus requiring neither the memory circuit for storing thedefective address nor the address comparator. This makes it possible tohighly integrate and reduce the power consumption of the semiconductormemory device. Since none of the aforementioned address comparison isrequired, the memory access can be speeded up.

In case, on the other hand, the aforementioned redundancy columnselection line is provided for a plurality of column selection lines, itis steadily supplied, if not used, with the ground potential toestablish the shield action for suppressing the coupling of the columnselection lines.

FIGS. 84(A) to 84(C) are waveform charts of one embodiment forexplaining the testing method of the word lines and a circuit diagramcorresponding to the former.

This embodiment is additionally provided with a control signal EM. Thissignal EM is newly added, as one test mode composed of address signalsin the aforementioned test mode, to that fed from the externalterminals. In the same Figure (A), there is shown a timing chart for theschematic selections of the word lines in the normal mode. In thisnormal mode, in accordance with the selections of the RAS system, theaddress designations A0 to A3 inputted select the corresponding wordlines sequentially.

In the aging mode (to be set as one of the test modes) having the signalEM at the high level, on the contrary, the selected word line WL1 ismaintained at the high level even if the RAS signal is reset from thelow level to the high level. If, therefore, the addresses A0 to A3advanced stepwise by the RAS signal are inputted, the word lines WL1 toWL3 selected sequentially as above are not reset irrespective of thehigh level of the RAS signal. These selected word lines WL1 to WL3 arereset by dropping the signal EM to the low level, although notespecially limitative.

In the same Figure (C), there is shown a circuit diagram of oneembodiment of the word line selection circuit. The signal EM has itslevel converted to the low level, if in the aging mode, by the levelconverter circuit which is constructed of a NOR gate circuit of latchmode using the boost voltage VCH as its operation voltage. As a result,the P-channel MOSFET is turned ON, and the P-channel MOSFET madereceptive of the word line reset signal WPHL and the P-channel MOSFETconnected therewith in series are turned OFF to invalidate the output ofthe P-channel MOSFET made receptive of the word line reset signal WPHL.As a result, the word line WL are held at the high level once they areraised to the high state.

When the word line WL is reset or when in the normal mode, theaforementioned level conversion output is raised to the high level (VCH)in response to the low level of the signal EM. As a result, theaforementioned P-channel is turned OFF, and the P-channel MOSFET madereceptive of the aforementioned signal WPHL and the P-channel MOSFETconnected therewith in series are turned ON so that the input of theCMOS inverter circuit for driving the word line WL is raised to the highlevel to reset the word line WL from the high to low levels.

Incidentally, at the input of the CMOS inverter circuit for driving theword line, there is provided a switch MOSFET which is to be controlledby the inverter circuit made receptive of the output signal. As aresult, when in the aforementioned multi-selection, the high level of anon-selected signal X0UB is prevented from being transmitted to the CMOSinverter circuit to maintain the aforementioned selection level.

In the aging, If the word lines are selected one by one with the signalEM being set at the high level, they can be held in the selected state.Since the high-level period of the selected word lines can be elongated,the stress duty can be raised to effect the efficient aging for arelatively short time period.

FIGS. 85(A) to 85(D) show one embodiment of the signal mount margin testmethod. In this example, a control signal SM is additionally provided.This signal SM is added not only from the external terminal but also asone test mode of the combination of the address signal in theaforementioned test mode. In the same Figure (A), there isrepresentatively shown each circuit such as the sense amplifier relatingto a pair of complementary bit lines, the precharge circuit, the columnswitch and the shared switch circuit.

In the same Figure (B), there is shown a waveform chart of theoperations of the normal mode. In this normal mode, the signal SM isdropped to the low level. In accordance with this, the shared selectionsignal SHL at the side of the selected word line (L), but the sharedselection signal SHR at the side of the unselected word line (R) isdropped to the unselected low level. Therefore, the stored memory isread out from the selected memory cell to the complementary bit line BL.

In the same Figure (C), there is shown a waveform chart of the signalamount test mode. In this test mode, the signal SM is raised to the highlevel. In accordance with this, not only the shared selection signal SHLat the side of the selected word line (L) but also the shared selectionsignal SHR at the side of the unselected word line (R) is raised to thehigh level. Thus, the righthand and lefthand bit lines BL are coupled tothe input of the sense amplifier so that the bit line capacity issubstantially doubled. Therefore, the read level of the storedinformation from the selected memory cell is dropped to about one halfin the aforementioned normal mode. In response to this, it is possibleto accomplish the signal amount margin test for testing whether or notthe sense amplifier performs its amplification accurately.

In the same Figure (D), there are presented circuit diagrams showing oneembodiment of the shared selection signal generator. The control signalSH is added to control the validity/invalidity of selection signals SLand SR through the NOR gate circuit. When the signal SM is at the highlevel, more specifically, both the signals SL/SR are forced to theselection level to raise the aforementioned signals SHL and SHR to thehigh selection level. Incidentally, this selection level is at the boostvoltage VCH like before.

FIG. 86 shows another embodiment of the function mode.

In response to the function set signal formed by the WCBR, binarynumerical data are inputted directly from the address terminals A0 toA3. These numerical data are converted into analog signals S0V to S1V bya voltage decoder (i.e., digital/analog converter), for example. Thisanalog voltage SiV is fed to the internal voltage generator constructedof the operation amplifier of voltage follower structure to form theinternal voltage VCC or VDL. This structure can set the internaloperation voltage at will. As a result, the voltage margin test or theacceleration test in the aging can be simplified.

On the other hand, the binary numerical data are inputted directly fromthe aforementioned address terminals A0 to A3 to the time decoder toform decode signals S0D to A10D, which signal SiD is inputted to a delaycircuit. This delay circuit has its delay time made variable from 0 to10 ns in accordance with the signals S0D to S10D. As a result, anarbitrary delay time can be attained in accordance with the signal SiD.This delay circuit is used as that for forming the time-series timingsignals of the RAS or CAS system, for example. By making use of thedelay circuit, it is possible to test the time margin, for example.

FIG. 87 shows another embodiment of the refresh address counter. To thisembodiment, there is added a control signal CS. This signal CS is fedfrom the external terminal, added as one test mode composed of thecombination of the address signals in the aforementioned test mode, orformed by the power-on detection signal.

In the same Figure (A), there is shown a waveform chart of the normalmode. In this normal mode, the signal CS is set at the low level. Inaccordance with this, the counter circuit accomplishes its countingoperation using the RAS signal as the clock to form a refresh addresssignal ARi when in the CBR refresh. In the same Figure (B), there isshown a waveform chart of the operations of the counter set. In thecounter set, the signal CS is raised to the high level. If, at thistime, the CBR is accomplished, the address signal to be inputted insynchronism with the low level of the RAS signal is inputted as theinitial value of the counter. When the signal CS takes the low level,the counter circuit holds its initial value at +1.

In the same Figure (C), there is shown the circuit diagram. In order tomake the aforementioned external input possible, there is added anexternal set input circuit which is controlled by the signal CS.

FIG. 88 shows another embodiment of the internal power monitor system.

In the same Figure (A), there is shown a block of this embodiment. Thevoltage VCC or VDL to be formed by the internal drop voltage powercircuit VCC or VDL is inputted to one input of the level comparator. Thereference voltage is fed through an external pin to the other input ofthe level comparator. This level comparator outputs the magnituderelation of the two as a binary signal to the external terminal DOUT.

In the same Figure (B), there is shown a waveform chart for explainingthe operations of this embodiment. As indicated by broken lines, thelevel of the voltage VDL can be obtained indirectly from the changingpoint of the high/low levels of the output signal DOUT by changing thevoltage to be fed to the external pin. The input voltage to be fed fromthe external pin may be either fed one-to-one to the level comparator orhave their level attenuated or amplified. Likewise, the voltage VCC orVDL may have its level attenuated at a constant ratio. In case the levelis thus attenuated, the level of the aforementioned boost voltage can bemonitored. The structure, in which the level comparator is disposedinside like this embodiment, is kept away from the influences of thelevel fluctuations in the output voltage pass unlike the system, inwhich the analog voltage is outputted as it is to the outside, so thatthe level can be monitored highly precisely.

FIG. 90 is a schematic section showing the element structure of oneembodiment of a memory cell portion, an N-channel type column switchMOSFET for the Y-selection and a P-channel MOSFET to be used in anotherCMOS circuit. In the same Figure, there is shown a section of theschematic element structure taken in the direction of bit lines.

The memory cell and the N-channel MOSFET constituting the column switchare formed in a P-type WELL formed over a P-type substrate 41.

As shown one pair of memory cells are provided for bit lines 50 made ofpolysilicide. For the source and drain 44 shared between the addressselection MOSFET constituting the paired memory cells, there is provideda pad contact 47 which is made or conductive polysilicon and formed inthe contact hole formed by the self-alignment technology. Across theshared source and drain 44, there are formed the source and drain 44 ofthe capacitor side. A gate electrode 46 is formed through a thin gateinsulating film 53 between the two regions. The gate electrode 46 ismade of conductive polysilicon and constitutes a word line. This wordlihe accomplishes the word shunt by an aluminum layer 52 formedthereover. In the same Figure, there is shown by way of example the wordline 46 which is connected with the gate of the address selection MOSFETof another memory cell which has its pitch offset vertically. The wordline 46 is formed over the relatively thick field insulating film.

The capacitor-side source and drain of the address selecting MOSFET areconnected with the conductive polysilicon 48 constituting the store nodeof the information storing capacitor. This polysilicon 48 is equippedthrough a thin insulating film 54 with the polysilicon 49 constitutingthe plate electrode of the aforementioned capacitor.

Over the bit line 50, there is formed a tungsten layer 51 acting as afirst metal layer for forming a column selection line. Although notespecially limitative, the polysilicide 50 constituting theaforementioned bit line is connected through the shared selection switchMOSFET, although not shown, with the aforementioned tungsten layer 51and further with the source and drain 44 of one of the MOSFETconstituting the columns switch of the same Figure. The I/O-side sourceand drain 44 are connected with the input/output lines I/O made of thesecond aluminum layer 52 through the first metal layer 51 through thepad contact 47 like the aforementioned address selecting MOSFET of thememory cell. Incidentally, the righthand side of the same Figure isprovided with a P-channel MOSFET. This P-channel MOSFET is used as asense amplifier or another CMOS. Thus, the P-channel MOSFET is formed inan N-type WELL 43 and composed of a source drain 45 and the gate 46.

In this embodiment, the N-channel MOSFET constituting the column switchto be connected with the aforementioned input/output lines I/O isexemplified by using the pad contact 47 similar to that of the addressselection MOSFETs of the memory cell as the source and drain to beconnected with the input/output lines I/O. In this structure, theself-alignment technology is used for boring the contact holes in theoxide film of the surface of the source and drain. As a result, thesource and drain below the pad contact 47 need not be made large whileconsidering the contact boring mask misalignment, so that the can beminimized. This makes it possible to increase the integration and toreduce the parasitic capacity. In case the sources and drains of thenumerous column switch MOSFETs like the input/output lines I/O areconnected, the paratisic capacity can be drastically reduced inaccordance with the reduction in the parasitic capacity of the sourceand drain of the column switch MOSFETs. As a result, the wiring capacityof the input/output lines I/O can thus be drastically reduced, thewrite/read operations can be speeded up.

The MOSFET using the aforementioned pad contact can be utilized in eachcircuit requiring the fine structure and the low parasitic capacity suchas not only the aforementioned column switch MOSFET but also a MOSFETconstituting the sense amplifier, a bit line precharging MOSFET, a bitline shorting MOSFET, a shared sense amplifier selecting MOSFET or aword line driver MOSFET.

FIG. 92 is a schematic circuit diagram showing another embodiment of themain amplifier selection circuit.

In the embodiment of the same Figure, the main amplifier MA is usedcommonly for memory mats which are divided vertically with respect tothe main amplifier MA. In other words, the main amplifier MA is arrangedat the center between a pair of memory mats each composed of a memorycell array M and a sense amplifier S. The input/output lines I/O andI/OB of the memory mats are selectively connected with the inputs of themain amplifier MA through a switch MOSFET which is to be switched by matselection signals MSU and MSD. The layout relation of the memory matsand the sense amplifiers is basically similar to that of the foregoingembodiment of FIG. 2 so that the number of the main amplifiers can bereduced.

If the number of main amplifiers is to be merely reduced, the mainamplifier MA can be arranged above the upper memory mat or below thelower memory mat. In this case, however, those of the wiring lines ofthe input/output lines to be connected with the input terminals of themain amplifier MA, which correspond to the memory mat at the oppositeside, are elongated. In the structure in which the main amplifier isarranged at the center of the divided memory mats, as in the embodimentsshown in the same Figure or foregoing FIG. 2, on the contrary, theinput/output lines I/O and I/OB to be arranged in the two memory matscan be equally shortened to speed up the memory accesses.

FIG. 93 is a schematic circuit diagram showing still another embodimentof the main amplifier selection circuit.

In the embodiment of the same Figure, the main amplifier MA is usedcommonly for the memory mats which are vertically divided and arrangedwith respect to the main amplifier MA. The memory mat of this embodimentis exemplified by the share sense amplifiers which have the memory cellarrays bisected to the right and left from the sense amplifier S. Inthis structure, the vertically divided memory cell arrays are deemed asthe memory mats and arranged with input/output lines I/O and I/OB sothat they are selectively connected with the input of the main amplifierMA through the switch MOSFET which are to be switched by mat selectionsignals MS0 to MS3. The layout relation between the memory mats and thesense amplifier is basically similar to that of the aforementionedembodiment of FIG. 2 so that it can reduce the number of the mainamplifiers and substantially shorten the length of the input/outputlines. In the structure, moreover, in which the input/output lines I/Oand I/OB are arranged for each pair of memory cell arrays M as in thisembodiment, the number of the column switch MOSFETs to be connected withthe input/output lines I/O and I/OB can be halved. As a result, thelength of the input/output lines can be substantially shortened, and thewiring capacity can be reduced so that the operations can be speeded up.

FIG. 94 is a layout showing another embodiment of the DRAM according tothe present invention.

This embodiment is based upon the foregoing layout of FIG. 2 andmodified such that the semiconductor chip is bisected by thelongitudinal center line and such that the layouts of FIG. 2 arearranged symmetrically with respect to that center line. In thisstructure, each of the halves of the memory chip is formed with crossareas composed of the aforementioned longitudinal and transverse centerareas. In case the memory chip is divided by the longitudinal centerline, as shown, the transverse center portions are arranged on one line.Thus, the memory arrays are divided into eight by the aforementioned twocross areas. Moreover, these two cross areas are arranged with theperipheral circuits and the bonding pads like the foregoing embodimentand are bonded by the LOC leads.

In case this layout is applied to the dynamic RAM having theaforementioned storage capacity such as 16 Mbits, the word line lengthcan be halved to accelerate the high-speed access. Moreover, the memorymats is more finely divided so that the power consumption can be furtherdropped. Furthermore, the aforementioned cross area and the four dividedareas are used as the fundamental components and arranged in two sets,as above, so that the storage capacity of the RAM can be furtherincreased.

In addition to the shown structure in which the memory chip is bisectedby the longitudinal center line and formed with the aforementioned crossareas, there may be another structure in which the memory chip isbisected by the transverse center line and formed with cross areas madeby a similar method. Moreover, these structure may be combined anddivided in another method.

FIG. 95 is a pattern diagram showing one embodiment of the memory cellarray according to the present invention.

The bit lines are crossed at a constant spacing to reduce the couplingnoises between the adjoining bit lines. If this bit line crossing isadopted, there arises a problem that the area of the bit line crossportions is increased. In this embodiment, therefore, the wiring layerto be used as the column selection line is used as the cross wiringline. In case the first metal layer is used as the column selectionline, as shown in the same Figure, there is used, for the bit line madeof a polysilicide layer to be replaced, the first metal wiring layer tobe formed thereover.

By adopting the structure using this first metal layer, the wiring layerespecially for the bit line cross portions can be eliminated.

In order to equalize the parasitic capacities of the aforementioned bitlines and the column selection lines extending in parallel, the columnselection lines are so folded at the bit line cross portion that areshifted by one pitch. As a result, one column selection line is enabledto have an equal parasitic capacity for the two-paired bit lines, andthe folded portions can be used as the bit line cross portions. Nospecial area is required for the bit line cross portions so that thecontinuity of the various wiring patterns is not deteriorated.

In case, on the other hand, the cross portion of the bit lines is formedby making use of the upper wiring layer, the uniformity of the capacityconstituting the underlying memory cell and the address selecting MOSFETis not adversely affected. This makes it possible to hold the continuityand uniformity of the devices (such as the capacitors or MOSFETs)constituting the memory cell and to reduce the dispersion of thecharacteristic margins of the individual bit lines. Since, moreover, thepattern continuity of the pattern is held and since the crossing contactis taken separately of the bit line contact, no problem raises in thefabricating and treating conditions.

This will be easily understood from a section shown in FIG. 96(A) and aschematic diagram shown in the same Figure (B). As shown in the sectionof the same Figure (A), at the cross portion of the bit lines, theunderlying bit line pairs of polysilicide are separated from each othersuch that one bit line is replaced, while being made of thepolysilicide, by the position of the other bit line whereas the otherbit line is intersected with the one bit line by the first metal layerformed thereover and is replaced by the position of one bit line.

FIGS. 97 to 99 are layouts showing one embodiment of a shared senseamplifier column and a corresponding memory cell array portion.

In FIG. 97, there are formed between the memory cell array portion atthe righthand side and the shared MOSFET dummy layers 69 and 70 whichconstitute a step damping region extending in the longitudinaldirection. In case the stacked memory cell is used as in thisembodiment, the step damping region has its memory cell array raised byabout 1 microns from the remaining peripheral circuit. As a result, thestep between the memory cell array portion and the peripheral circuitportion becomes so steep as to make it difficult to work the wiringlayers and to open the contact holes in the vicinity of the step.

As shown in the same Figure, the first polysilicon layer 69 to be formedsimultaneously with the gate electrode of the MOSFET and the stepdamping word line 70 are formed as dummy layers. In this structure, asis apparent from the section of FIG. 100, the step between the memorycell array portion and the peripheral circuit portion can be made gentleby forming the aforementioned dummy layers.

In this embodiment, moreover, the step damping region is utilized andformed with an N⁺-type diffusion layer so that the guard ring functionof the memory cell array portion is given by supplying the voltage VDL.As a result, the minority carriers, which are generated by theoperations at the peripheral circuit side, can be prevented fromreaching the memory cell array portions and being coupled to the storedcharges so that the holding time period is not shortened.

FIG. 98 is a pattern diagram showing the P-channel MOSFET constituting aY-gate (i.e., the column switch MOSFET) arranged at the lefthand side ofFIG. 97 and the sense amplifier. FIG. 99 is a pattern diagram showing abit line precharge MOSFET arranged at a lefthand side, an N-channelMOSFET constituting the sense amplifier, a shared MOSFET and a memorycell array at the lefthand side. Thus, the step,damping region is formedbetween the lefthand memory cell array portion and the shared MOSFET.

In FIGS. 97 to 99, reference numeral 61 designates a bit line which ismade of polysilicide and arranged to extend in the transverse direction.Numeral 62 designates a column selection which is made of a first metallayer extending in the transverse direction like the foregoingembodiment. Numeral 63 designates a word line which is made of apolysilicon layer and shunted by a second metal layer 68 formedthereover. This word line is arranged to extend in the longitudinaldirection, as shown. Numeral 64 designates an address selection MOSFETconstituting a memory cell. A memory capacitor is omitted so as to avoidthe complicatedness of the pattern. Numeral 65 is a bit line contactwhich is formed with the pad contact like the foregoing embodiment.Numeral 66 designates a diffusion layer. Numeral 67 designates aninput/output line I/O which is made of a second metal layer like theword shunt and arranged to extend in the longitudinal direction, asshown in the same Figure.

By making use of the step damping region, moreover, the polysiliconconstituting the gate of the shared MOSFET is shunted to drop thesubstantial resistance thereby to form the second metal layer forspeeding up the operations.

FIGS. 101 to 108 are pattern diagrams showing one embodiment of thememory cell array portion in the direction of the word lines and thecorresponding peripheral circuit.

In FIG. 101, the aforementioned step damping region is formed at thelefthand side of the memory cell array. For this step damping, there isformed a dummy polysilicon wiring line 78. The substrate surface isformed below the step damping region with a guard ring diffusion layerof memory cell array and a wiring layer for applying the bias voltageVDL.

In the memory cell array, reference numeral 71 designates a diffusionlayer, and numeral 72 designates a word line made of a polysiliconlayer. From the same Figure, the pattern of the capacitor is omitted.Numeral 73 designates a bit line which is made of the aforementionedpolysilicide. Numeral 74 designates a second metal layer for wordshunting. Numeral 75 designates a column selection line which is made ofa first metal layer. Numeral 76 designates a bit line contact which ismade of the aforementioned pad contact.

At the lefthand side of the memory cell array portion, there is formedacross the step damping region a word driver. In this word driver,reference numeral 79 designates the gate of a word driver MOSFET, andnumeral 80 deisgnates the output side first metal layer which isconnected with the word line of the driver MOSFET. Numeral 81 deisgnatescontacts to be connected with the source and drain diffusion layers ofthe MOSFET. The whole word driver is arranged to extend leftward in theorder of FIGS. 102 to 105 from the lefthand side of FIG. 101.

At the lefthand side of the aforementioned word driver shown in FIG.105, there is arranged an X-decoder which extends leftward, as shown inFIGS. 106 and 107.

In FIG. 108, there is shown a pattern diagram of one embodiment of theword clear circuit disposed at the righthand side of the memory cellarray portion shown in FIG. 101, i.e., at the other end of the wordline, with which is connected the output of the word driver.

In the same Figure, too, there is formed between the righthand end ofthe memory cell array portion and the word clear circuit a step dampingregion which is similar to the aforementioned one. This region is formedwith a step damping wiring (polysilicon) and guard ring shunt 99.

In the same Figure, reference numeral 91 designates a word clear signalline which is formed of a second metal layer. Numeral 92 designates aground line which is formed of a first metal layer. Numeral 93designates a gate of the word clear, which is made of a polysiliconlayer. Numeral 94 designates a diffusion layer. Numeral 95 designates astep damping dummy polysilicon layer. Numeral 96 designates a word lineshunting layer which is formed of a second metal layer. Numeral 97designates a word line made of polysilicon. Numeral 100 designates a bitline made of polysilicide. On the other hand, black squares designatecontact portions.

The operational effects obtained from the embodiments thus far describedare as follows:

-   (1) The semiconductor chip is arranged with the peripheral circuits    in a cross area composed of the longitudinal center portions and the    transverse center portions, and memory arrays are arranged in the    four regions which are divided by the cross area. In this structure,    the longest signal transmission pass can be shortened to about one    half of the chip size in accordance with the arrangement of the chip    center portion with the peripheral circuit, so that the DRAM    intended to have a large storage capacity can be speeded up. If,    moreover, the cross area is formed for the two regions which are    bisected by the longitudinal center line of the semiconductor chip    thereby to adopt the aforementioned layout, there can be attained a    better effect that the storage capacity and the operating speed can    be further improved.-   (2) The edges of the cross area contacting with the memory arrays    are arranged with X-decoders and Y-decoders so that the signal    transmission passes to be formed in the cross area for the address    buffers and the predecoders can be shortened. As a result, there can    be effected a rational layout and a high speed.-   (3) The regions of the longitudinal or transverse center portion of    said cross area, which are interposed between the X-decoders, are    arranged with at least one of a main amplifier, a common source    switch circuit, a sense amplifier control signal generator and a mat    selection control circuit. As a result, the peripheral circuits    arranged in the cross area, which correspond to the X-decoders, the    sense amplifiers, and the input/output lines I/O, can be disposed in    their vicinities to rationalize the layouts of memory cell selection    circuits and the storage information transmission passes so that the    high integration and the high speed can be attained.-   (4) The regions of the longitudinal or transverse center portion of    said cross area, which are interposed between the Y-decoders, are    arranged with at least one of an address buffer, a control logic    circuit corresponding to a control signal, and a defect relief    circuit. According to this structure, a rational layout according to    the signal transmission pass can be realized to speed up the    operations.-   (5) The center portion of said cross area, where the longitudinal    and transverse center portions are superposed, is arranged with at    least one of at least the final driver circuit of a decoder    inputting address signal generator and a power generator to be used    inside. As a result, for the X- and Y-decoders for selecting the    word lines and column selection lines, their input signals are    transmitted four ways from the center of the chip. Thus, the signal    transmission passes are divided and shortened, and the loads are    shared and lightened so that the high-speed operations can be    realized.-   (6) Those circuits of the peripheral circuits, which may probably    inject minority carriers into a substrate on principle, are arranged    on two center lines of the cross area or their vicinities. While    retaining the high-speed operations by arranging the peripheral    circuits at the center of the chip, the influences of the minority    carriers upon the memory cell array portions can be minimized.-   (7) The memory arrays formed in the four quartered areas of said    cross area are constructed of a block of plural memory mats at a    unit having the same size as includes the sense amplifiers. Thanks    to this structure, the mat selecting operations by the addresses of    higher order are added to the selecting operations of the memory    cells in the mat so that the selecting operations can be divided    into two stages so that the decoder can be accordingly divided. As a    result, there can be attained an effect that the loads upon the    decode signal can be lightened to speed up the operations.-   (8) The memory arrays formed in the four quartered areas of said    cross area are arranged with at least X-decoders or Y-decoders in a    manner to divide each of the memory arrays. As a result, the word    lines or column selection lines can be shortened accordingly as they    are substantially divided by the decoders, so that the memory cells    can be selected at a high speed.-   (9) Each unit memory mat includes a control circuit for generating a    variety of timing signals for the memory cell selections on the    basis of a mat selection signal. As a result, the time-series    operation sequence can be executed in the memory mat with the    optimized timing. In the DRAM of large storage capacity composed of    a number of memory blocks, the timing margin between the different    memory blocks need not be taken so that the fast memory access and    the increased operation margin can be attained. Moreover, the number    of operating memory mats can be easily changed to facilitate the    development of kinds (i.e., to drop the power).-   (10) Two adjacent ones of the unit memory mats are paired to form    one sub-block which is equipped with a control circuit for    controlling the memory mats. In this structure, one memory mat can    be selected in the sub-block so that the control circuit can be used    commonly for the plural memory mats to increase the high integration    and speed up the operations.-   (11) A pair of axially symmetric sub-blocks are used to construct    said unit memory mat. As a result, the control circuit can be used    commonly for more memory mats to increase the high integration and    speed up the operations.-   (12) The control circuit is activated by the mat selection signal,    the sub-block selection signal or the block selection signal. As a    result, the wasteful current consumption at the unselected mat or    sub-block can be suppressed to drop the power consumption.-   (13) The control circuit controls at least one of the precharge of    complementary data lines, the activation of sense amplifiers, the    control of shared sense amplifiers, the activation of X-decoders,    the activation of Y-decoders, the activation of word drivers, the    selection of common input/output lines, the selection of main    amplifiers, and the activation of main amplifiers. As a result, the    operation sequence control in the mat can be optimized.-   (14) The unit memory mats are fed with selection signals for    selecting word lines and complementary data lines belonging thereto.    In this structure, the selection signals are formed by the    pre-decode circuit so that the decoder circuit can be rationally    divided.-   (15) The circuits for generating the selection signals for selecting    the word lines or complementary data lines belonging to said unit    memory mats are provided commonly for the plural memory mats or    sub-blocks. As a result, the excess handling of the mat control    signals can be eliminated to drop the power and speed up the    operations.-   (16) The selection signals of said memory mats or memory blocks are    generated by decoding the address signals inputted through an    address buffer especially therefor. Thanks to this structure, the    address signal for forming the mat selection signals can be    separated from the relatively large load capacities such as the    numerous address comparators disposed in the redundancy circuit, so    that the operation can be speeded up and so that the mat selections    can be accomplished prior to the selections of the memory cell    arrays.-   (17) Said cross area is arranged in its region with a portion or all    of bonding pads. As a result, the signals can be exchanged from the    center portion of the chip the signal transmission passes are    extended four ways from the chip center portion to the periphery.    Despite of the large size of the chip, however, the signal    transmission passes can be shortened to speed up the operations.-   (18) All of said bonding pads are zigzag arrayed in two rows in the    longitudinal center portion of said cross area. As a result, the    numerous bonding pads can be efficiently arranged for the high    integration.-   (19) The bonding pads arrayed in the longitudinal center portion of    said cross area are bonded to an LOC lead frame. As a result, this    lead frame can be formed into a portion of the wiring for the power    supply pad, and the bonding pads can be disposed in the vicinity of    the input circuit to improve the level margin and speed up the    operations.-   (20) Those of bonding pads, which are used for applying the power    voltage or the circuit and the ground potential, are arranged at a    suitable spacing according to circuit blocks requiring them and are    connected to the common LOC lead frame to be fed with the power    voltage of the circuit and the ground potential. As a result, the    level of the noises accompanying the circuit operations can be    suppressed to improve the operation margin.-   (21) Those of said bonding pads for applying the ground potential    are disposed in plurality along the chip distribution of the sense    amplifier rows to be activated. As a result, the relatively high    currents by the amplifications of the sense amplifiers are fed from    the corresponding pads so that the level of the noises generated in    the potential voltage of another circuit can be suppressed at a low    level to enlarge the operation margin.-   (22) The cross area of a semiconductor chip composed of a    longitudinal center portion and a transverse center portion is    arranged with peripheral circuits and bonding pads, and the four    regions quartered by said cross area are arranged with memory    arrays. Moreover, said semiconductor chip has its four corners    stepped. As a result, the stress coming from the mold resin at the    corners of the chip can be prevented from being applied directly to    the memory cell portions.-   (23) The steps formed in the four corners of said semiconductor chip    are constructed by stacking the wiring layers which are formed by    the sane step as that for said memory arrays. The stress to be    applied to the chip from the mold resin can be dispersed without    adding the production step.-   (24) The cross area of a semiconductor chip composed of a    longitudinal center portion and a transverse center portion is    arranged with peripheral circuits. The four regions quartered by    said cross area area arranged with memory arrays. Said semiconductor    chip has its outermost periphery arranged with a highly doped    diffusion layer of the same conductivity type, which is to be fed    with a substrate back bias voltage. Said semiconductor chip further    has its inside arranged with a guard ring which is made of a    diffusion layer of the polarity opposite to that of said substrate    and which is to be fed with the power voltage. Thanks to this    structure, it is possible to prevent the undesired noises from    stealing into the memory array portions.-   (25) The semiconductor memory device is constructed to comprise an    internal drop voltage generator made operative in response to the    power voltage fed from an external terminal and including one or    more impedance converting output buffers made receptive of a    reference voltage prepared by a reference voltage generator. Thanks    to this structure, it is possible to drop the operation voltage    according to the drop of the breakdown voltage accompanying the    finer structure of the element and to drop the power consumption by    the drop of the operation voltage. Since, moreover, the drop voltage    is generated by the constant reference voltage, the operations of    the internal circuit can be stabilized because they are freed from    the influences of the fluctuations of the external power voltage.-   (26) Said internal drop voltage generator is provided for each of a    memory array operating voltage and a peripheral circuit operating    voltage. As a result, it is possible to prevent the noises from    being generated by the circuit operations.-   (27) The drop voltage to be generated by said internal drop voltage    generator is set at a level twice as high as the logic threshold    voltage of an input buffer circuit to be fed therewith. As a result,    the operation voltage can be effectively utilized to enlarge the    input level margin.-   (28) The output buffer for said impedance conversion has an output    circuit made of the CMOS structure and is given a function to output    the power voltage selectively through the power voltage side    P-channel one of said output MOSFETS. Without any circuit being    added, therefore, there can be attained a function to switch the    internal operation voltage to the power voltage fed from the    outside. This voltage switching function can be utilized for the    aging.-   (29) An internal drop voltage generator is made operative in    response to a power voltage fed from an external terminal for    generating an operating voltage of an internal circuit, and a level    conversion circuit converts the output signal formed by said    internal circuit to a signal level corresponding to the power    voltage fed from said external terminal. The output MOSFET of the    source follower type has its gate fed with the signal to be    outputted through said level conversion circuit. In this structure,    not only the level amplitude of the output signal but also the    amplitude of the drive signal can be increased to speed up the    operations.-   (30) The output circuit including the output MOSFET of the source    follower type is constructed such that the output MOSFET, which is    made receptive of the signal formed by said internal circuit as it    is, is disposed in parallel with the output MOSFET made receptive of    the signal through said level conversion circuit. As a result, the    conversion of the output signal can be started at a relatively quick    timing so that the signal conversion can be linearly accomplished    for a relatively long time. Thus, it is possible to drop the level    of the noises made on the power line or ground line when the output    signal is converted, without sacrificing the output operation speed.-   (31) The drop voltage generated by the internal drop voltage    generator is selectively outputted, in the output high impedance    state of a data output buffer in a test mode, from the output    terminal of the data output buffer through a switch MOSFET to be    switched by a signal at the bootstrap voltage or external power    voltage level. As a result, whether or not the internal power    circuit is normally operating can be monitored to retain the high    reliability.-   (32) The selection signal of word lines or shared sense amplifiers    is formed by a selection circuit which is to be operated by a high    voltage generated by boosting the internal drop voltage. As a    result, the boost voltage can be stabilized without being influenced    by the external power, and the selections of the word lines can be    speeded up.-   (33) The memory cell arrays are arranged symmetrically with respect    to the main amplifier, and the main amplifier is selectively    connected with the input/output lines of the paired memory cell    arrays through a switch MOSFET to be switched in accordance with the    selections of the paired memory cell arrays. Thanks to this    structure, the number of the main amplifiers can be reduced, and the    substantial wiring length of the input/output lines can be shortened    to speed up the operations.-   (34) Said memory cell arrays adopt shared sense amplifiers having    sense amplifiers arranged at the center portion of the pair of its    bisected data lines, and four pairs of input/output lines    corresponding to the data line pairs divided by said sense    amplifiers are connected with said main amplifiers through the    switch MOSFET which is to be switched to correspond to the    selections of said memory cell arrays. In this structure, the data    line length of the shared sense amplifiers can be shortened, and the    input/output lines can be accordingly divided to halve the wiring    capacity of the input/output lines so that the operations can be    speeded up.-   (35) Said memory cell arrays are said unit memory mats. As a result,    the number of the main amplifiers can be reduced, and the wiring    length of the input/output lines coupled thereto can be shortened to    realize the high-speed operations.-   (36) The latch circuit is provided for latching a selection signal    for word lines in response to a control signal so that a word line    drive signal may be formed by the output signal of said latch    circuit. As a result, the word lines can be sequentially selected in    a multiplex manner so that the aging can be efficiently    accomplished.-   (37) Shared sense amplifiers are given an operation mode for    connecting both selected/unselected data lines. As a result, the    signal amount from the memory cell is reduced to one half as the    capacity of the complementary data lines is doubled, so that the    margin test of the signal amount can be easily executed.-   (38) There is provided in a function setting mode a function to set    the state of an internal circuit at a voltage or delay time    corresponding to a digital signal which has a plurality of bits and    which is inputted from an address terminal of plural bits. As a    result, the internal operation voltage and the signal delay can be    easily changed to accomplish the internal test efficiently.-   (39) There is provided a refresh address counter circuit which has    an additional resetting or initial value setting function in    response to predetermined external control signal. As a result, the    refresh operations can be utilized for address selections of the    multiple selections and the various read/write testing address    selections.-   (40) An internal power voltage generator is provided for forming a    voltage for operating an internal circuit and having a power monitor    function to compare a voltage based upon the internal voltage and a    voltage given from the outside to output a binary signal of the    compared result. Thanks to this structure, the internal operation    voltage can be monitored in high precision.-   (41) At least one of the pull-up MOSFET of CMOS structure composed    of the sense amplifier, the initial-step circuit of the main    amplifier and the input/output lines, the short MOSFET composed of    the complementary data lines and the complementary input/output    lines, and the MOSFET of diode mode constituting the charge pump    circuit is caused to have a low threshold voltage. As a result, the    operations can be speeded up.-   (42) At least one of a column switch MOSFET, a MOSFET constituting a    sense amplifier, a precharge MOSFET, a shorting MOSFET, a word line    driving MOSFET, and a MOSFET for cutting shared sense amplifiers has    its source and drain contact made of a pad contact similar to the    source and drain contact of a memory cell address selecting MOSFET.    As a result, like the memory cell, the self-alignment technology can    be utilized as the source-drain contacts so that these source-drain    regions can be minimized, as necessary. Thus, the high integration    and the low parasitic capacity can be achieved to speed up the    operations.-   (43) The cross portion of the bit line cross type makes use of the    first metal wiring layer formed over the wiring layer forming the    bit lines so that the wiring lines constituting the cross portion    can be eliminated but that the equalities of the surfacing capacitor    and the MOSFET are not adversely affected.-   (44) The first metal wiring layer also forms the column selection    lines, one of which is formed to correspond two pairs of bit lines    and folded to overlap from one to other bit line pair at portion    different from the cross portion of the bit lines. Any special cross    wiring region is not required, but the parasitic capacity between    the column selection lines and the bit lines can be equalized.-   (45) A step damping region made of a dummy wiring layer is formed    between a memory cell array portion of laminated type and a    peripheral circuit. As a result, it is easy to treat the wiring    lines.-   (46) Said step damping region is formed therebelow with a guard ring    so that the characteristics can be stabilized.-   (47) There is constructed the memory array of a block which is    composed of plurality units of memory mats having the same size and    including sense amplifiers; forming redundancy word lines and/or    redundancy data lines for each of said memory mats. The redundancy    decoders of a number smaller than the total number of the redundancy    word and/or data lines of all of said memory mats and larger than    the total number of the redundancy word and/or data lines of each of    said memory mats are formed so that they may be used for each of    said memory mats or commonly for said memory mats. As a result, the    circuit scale necessary for defect remedy can be reduced to raise    the integration and reduce the power consumption.-   (48) Said redundancy decoder includes a defect address memory    circuit and an address comparator and is arranged in proximity to    corresponding X- and Y-address buffers. As a result, the signal    transmission passes can be made the shortest to speed up the    operations and effect the high integration.-   (49) The preparatory word lines and/or preparatory column selection    lines wired to intersect a plurality of word lines and/or column    selection lines, respectively, are formed at the output of a word    line or column selector. The word lines and/or the output lines of    the column selector are cut by physical means, when a word line    and/or a data line are defective, from the column selection lines    corresponding to the defective word line and/or the defective data    line and are connected with the preparatory word lines and/or the    preparatory column selection lines. In this structure, the memory    circuit and comparator circuit for the defective address can be    eliminated to increase the integration and drop the power    consumption.-   (50) When in the multi-bit simultaneous testing mode by the    multiplex selection of the column system, only the defective data or    column selection line of the data lines or column selection lines is    switched to a redundancy data line or a redundancy column selection    in a manner to correspond to the memory cell array divided into a    plurality of memory blocks. As a result, the number of the    redundancy data lines or YS lines to be prepared can be reduced    while shortening the test time resorting to the aforementioned    multi-bit simultaneous testing function.-   (51) The data lines are divided into a plurality of blocks by one of    a specific-bit of the address signals of the row and/or column    systems, a block address prepared inside, or the combination of the    address signal and the block address, and a defective data line in a    defective block only is switched to a redundancy data line by making    use of a signal designating the block. Thus, the number of the    redundancy data lines or YS lines to be prepared can be reduced.-   (52) The word lines are divided into a plurality of blocks by    assigning block addresses formed in the row system and/or inside, so    that the defective word line may be switched to a redundancy word    line only in the block having the defective word line by making use    of a signal designating said block. Thus, the number of the    redundancy word lines to be prepared can be reduced.-   (53) Said block address is designated by the same program means as    that for programming the defective address so that the program can    be simplified.

Although our invention has been specifically described hereinbefore inconnection with the embodiments thereof, it should not be limited tothose embodiments but could naturally be modified in various mannerswithout departing from the gist thereof. For example, the storagecapacity of the dynamic RAM should not be limited to the aforementionedvalue of 16 Mbits but might be 4 Mbits or less or 64 Mbits or more.Moreover, there may be used the non-multi system, in which the X- andY-addresses are fed as the address inputs from independent terminals, sothat the storage capacity may be accordingly at about 8 Mbits or 24Mbits.

The present invention can be applied widely to the semiconductor memorydevice having the aforementioned large storage capacity.

The effects obtainable from the representative of the inventiondisclosed herein will be briefly described in the following.Specifically, the semiconductor chip is arranged with the peripheralcircuits in a cross area composed of the longitudinal center portionsand the transverse center portions, and memory arrays are arranged inthe four regions which are divided by the cross area. In this structure,the longest signal transmission pass can be shortened to about one halfof the chip size in accordance with the arrangement of the chip centerportion with the peripheral circuit, so that the DRAM intended to have alarge storage capacity can be speeded up. The memory arrays formed inthe four quartered areas of said cross area are constructed of a blockof plural memory mats at a unit having the same size as includes thesense amplifiers. Thanks to this structure, the mat selecting operationsby the addresses of higher order are added to the selecting operationsof the memory cells in the mat so that the selecting operations can bedivided into two stages so that the deocoder can be accordingly divided.As a result, there can be attained an effect that the loads upon thedecode signal can be lightened to speed up the operations. Each unitmemory mat includes a control circuit for generating a variety of timingsignals for the memory cell selections on the basis of a mat selectionsignal. As a result, the time-series operation sequence control can beexecuted in the memory mat with the optimized timing so that the fastmemory access and the increased operation margin can be attained.Moreover, the number of operating memory mats can be easily changed tofacilitate the development of kinds. All of said bonding pads are zigzagarrayed in two rows in the longitudinal center portion of said crossarea. Thus, the numerous bonding pads can be efficiently arranged andare bonded to an LOC lead frame. As a result, this lead frame can beformed into a portion of the wiring for the power supply pad, and thebonding pads can be disposed in the vicinity of the input circuit toimprove the level margin and speed up the operations. The cross area ofa semiconductor chip composed of a longitudinal center portion and atransverse center portion is arranged with peripheral circuits andbonding pads, and the four regions quartered by said cross area arearranged with memory arrays. At the four corners of the for regions,there are stacked the wiring layers which are formed by the same step asthat for the memory array portions, so that the stress coming from themold resin of the chip can be dispersed. The semiconductor memory deviceis constructed to comprise an internal drop voltage generator madeoperative in response to the power voltage fed from an external terminaland including one or more impedance converting output buffers madereceptive of a reference voltage prepared by a reference voltagegenerator. Thanks to this structure, it is possible to drop theoperation voltage according to the drop of the breakdown voltageaccompanying the finer structure of the element and to drop the powerconsumption by the drop of the operation voltage. Since, moreover, thedrop voltage is generated by the constant reference voltage, theoperations of the internal circuit can be stabilized because they arefreed from the influences of the fluctuations of the external powervoltage. Said internal drop voltage generator is provided for each of amemory array operating voltage and a peripheral circuit operatingvoltage. As a result, it is possible to prevent the noises from beinggenerated by the circuit operations. The drop voltage generated by theinternal drop voltage generator is selectively outputted, in the outputhigh impedance state of a data output buffer in a test mode, from theoutput terminal of the data output buffer through a switch MOSFET to beswitched by a signal at the bootstrap voltage or external power voltagelevel. As a result, whether or not the internal power circuit isnormally operating can be monitored to retain the high reliability. Theselection signal of word lines or shared sense amplifiers is formed by aselection circuit which is to be operated by a high voltage generated byboosting the internal drop voltage. As a result, the boost voltage canbe stabilized without being influenced by the external power, and theselections of the word lines can be speeded up. At least one of thepull-up MOSFET of CMOS structure composed of the sense amplifier, theinitial-step circuit of the main amplifier and the input/output lines,the short MOSFET composed of the complementary data lines and thecomplementary input/output lines, and the MOSFET of diode modeconstituting the charge pump circuit is caused to have a low thresholdvoltage. As a result, the operations can be speeded up. At least one ofa column switch MOSFET, a MOSFET constituting a sense amplifier, aprecharge MOSFET, a shorting MOSFET, a word line driving MOSFET, and aMOSFET for cutting shared sense amplifiers has its source and draincontact made of a pad contact similar to the source and drain contact ofa memory cell address selecting MOSFET. As a result, like the memorycell, the self-alignment technology can be utilized as the source-draincontacts so that these source-drain regions can be minimized, asnecessary. Thus, the high integration and the low parasitic capacity canbe achieved to speed up the operations. The cross portion of the bitline cross type makes use of the first metal wiring layer formed overthe wiring layer forming the bit lines so that the wiring linesconstituting the cross portion can be eliminated but that the equalitiesof the surfacing capacitor and the MOSFET are not adversely affected.The first metal wiring layer also forms the column selection lines, oneof which is formed to correspond two pairs of bit lines and folded tooverlap from one to other bit line pair at portion different from thecross portion of the bit lines. Any special cross wiring region is notrequired, but the parasitic capacity between the column selection linesand the bit lines can be equalized. A step damping region made of adummy wiring layer is formed between a memory cell array portion oflaminated type and a peripheral circuit. As a result, it is easy totreat the wiring lines.

There is constructed the memory array of a block which is composed ofplural units of memory mats having the same size and including senseamplifiers; forming redundancy word lines and/or redundancy data linesfor each of said memory mats. The redundancy decoders of a numbersmaller than the total number of the redundancy word and/or data linesof all of said memory mats and larger than the total number of theredundancy word and/or data lines of each of said memory mats are formedso that they may be used for each of said memory mats or commonly forsaid memory mats. As a result, the circuit scale necessary for defectremedy can be reduced to raise the integration and reduce the powerconsumption. When in the multi-bit simultaneous testing mode by themultiplex selection of the Y-system or when the data lines or word linesare to be divided into a plurality of blocks by the address signal orthe internally formed block address or their combination, only thedefective block is replaced by the redundancy data line and theredundancy word line so that the number of the redundancy data lines orredundancy word lines to be prepared can be reduced.

Next, with reference to FIG. 109 to FIGS. 126(a) and 126(b), therefreshing operations of the DRAM, i.e., the refreshing system suitablefor a DRAM having a large capacity of 16 Mbits or more will be describedin the following in connection with the contents we have examined ordevised.

For the capacity of n bits, the DRAM of the prior art is constructed tohave refresh cycles in such a number of ½·√{square root over (n)} as isnecessary for refreshing the memory cells of n bits, and senseamplifiers in a number of 2·√{square root over (n)} to be activated inone refreshing operation.

The DRAM refresh is disclosed in Japanese Patent Laid-Open No.62-154291, for example.

The power to be consumed by the DRAM is occupied at a large ratio by thecurrent which is used to charge or discharge the bit lines through thesense amplifiers to be activated for refreshing the memory cell. Thiscurrent gets the higher for the larger capacity of the DRAM and occupiesthe larger ratio in the total consumed current. The rise in the powerconsumption increases the heat value of the chip to raise the chiptemperature. This rise in the chio temperature leads to the rise in thejunction temperature of the memory cell diffusion layer so that thejunction leakage is increased to deteriorate the information holdingcharacteristics of the memory cell.

In the DRAM of the prior art, for the capacity of n bits, the refreshcycle number is set at ½√{square root over (n)}, and the number of senseamplifiers to be simultaneously activated is set at 2√{square root over(n)}. In the system in which both the row and column addresses areinputted by the address multiplex, the number of sense amplifiers to beactivated for accessing predetermined memory cells may be √{square rootover (n)}. If a 2√{square root over (n)} number of sense amplifiers areactivated, there arises a problem that the current consumption may beaugmented. The present invention to be described hereinafter has anobject to reduce the current.

This object can be achieved by setting the number of sense amplifiers tobe simultaneously activated at √{square root over (n)} or less (e.g.,√{square root over (n)}, 1/2×·√{square root over (n)}, ¼·√{square rootover (n)}, ⅛·√{square root over (n)}, - - -, and so on). In other words,the number of refresh cycles is set at Vn or more (e.g., √{square rootover (n)}, 2√{square root over (n)}, 4√{square root over (n)}, 8√{squareroot over (n)}, - - - , and so on). However, such a large refresh cyclenumber will increase the number of refresh operations for refreshing allthe memory cells within a constant time (i.e., refresh interval) therebyto drop the memory efficiency of the DRAM. In order to prevent thememory efficiency from being dropped, it is necessary to elongate therefresh interval and accordingly to improve the information holdingcharacteristics of the memory cells. Since the information charges ofthe memory cells are reduced as a result of the junction leakage betweenthe substrate and the diffusion layer, the information holdingcharacteristics can be improved by using the memory cells ofstereoscopic structure, which can reduce the junction area but does notuses the substrate as the electrode of the information charge storagecapacity. Thus, the low power consumption can be achieved by increasingthe aforementioned refresh cycle number.

By setting the refresh cycle number at √{square root over (n)} or more,the number of sense amplifiers to be simultaneously activated can bereduced to √{square root over (n)} or less. As a result, it is possibleto reduce the bit line charging and discharging currents flowing throughthe sense amplifiers when in the refresh operation.

The present invention will be described in the following in connectionwith the embodiments thereof with reference to the accompanyingdrawings.

FIG. 110 shows the system of the prior art in which, for the capacity ofn bits, the refresh cycle number is ½·√{square root over (n)} whereasthe number of sense amplifiers to be simultaneously activated is2√{square root over (n)}. Two word lines are selected by the rowaddresses, and a 2√{square root over (n)} number of memory cells to beconnected therewith are refreshed. One of the two word lines is used notfor accessing the memory cells but is selected for the refreshing.

FIG. 109 shows one embodiment of the present invention, in which, forthe capacity of n bits, the refresh cycle number is √{square root over(n)} whereas the number of sense amplifiers to be simultaneouslyactivated is √{square root over (n)}. One word line is selected by therow address so that an √{square root over (n)} number of memory cellsconnected therewith are refreshed.

FIG. 112 shows the DRAM of ¼ n words×4 bits structure according to theprior art. Both the row address and the column address are halved sothat the two bits are selected from each of the two selected word linesto provide a structure of 4 bits. The refresh cycle number is ½·√{squareroot over (n)}. Each of the row and column addresses is reduced by one.

FIG. 111 shows the DRAM of ¼ n words×4 bits structure according to theprior art. The column address is quartered so that four bits areselected from one word line to provide a structure of 4 bits The refreshcycle number is √{square root over (n)}. Only the column addresses arereduced by two.

FIG. 114 shows the structure of a nibble mode of the prior art. Like thecase of the structure of 4 bits, two bits are selected from each of twowords so that they are accessed sequentially bit by bit by the toggle ofthe CAS. In this case, the addresses (i.e., nibble addresses) to besequentially selected internally are in each of the row and columndirections. The refresh cycle number is ½·√{square root over (n)}.

FIG. 113 shows the structure of a nibble mode according to the presentinvention. Like the case of the structure of 4 bits, four bits areselected from one word line so that it is accessed sequentially bit bybit by the toggle of the CAS. In this case, the nibble addresses are twocolumn addresses. The refresh cycle number is √{square root over (n)}.

FIG. 115(a) is a diagram comparing the address system of n words×1 bitstructure between the system of the prior art and the embodiment of thepresent invention. In the system of the prior art, the refresh addresses(i.e., the row addresses necessary for refreshing all the memory cells)have a number of (i−1) of the an i-number of row addresses excepting themost significant address. On the other hand, the nibble addresses arethe most significant row and column addresses. In the embodiment of thepresent invention, the refresh addresses are all of the i-number of rowaddresses, and the nibble addresses are the two more significant columnaddresses.

FIG. 115(b) shows the case comparing the address system of ¼ n words×4bit structure between the system of the prior art and the embodiment ofthe present invention. In the system of the prior art, both the row andcolumn addresses are deficient in the most significant one, as comparedwith the structure of x words×1 bit, and the refresh addresses are (i−1)in number like the structure of 1 bit. In the embodiment of the presentinvention, on the other hand, the column addresses are deficient in two,as compared with the structure of 1 bit, and the refresh addresses havean i-number like the structure of 1 bit. There are two address pinswhich do not belong to the address multiplex, so that the number of theaddress pins for the structure of ×1 is not reduced.

FIG. 116(a) shows an example of the external package view and the pinarrangement of the DRAM of 16 Mbits according to one embodiment of thepresent invention. FIG. 116(b) shows an example of the external packageview and the pin arrangement of the DRAM of 4 Mbits according to oneembodiment of the present invention. The shown case corresponds to n=16Mbits. The package has an external view of the small outline j-bentpackage SOJ, the zigzag line package ZiP, and the structures of 16Mwords×1 bit and 4 Mwords×4 bits. The difference from the system of theprior art resides in that the A11 pin is present in the structure of 4bits. Moreover, the A10 and A11 pins are not the address multiplex inputpins.

FIG. 117(a) shows the memory cell structure using the stacked capacitorSTC which is adopted in one embodiment of the present invention.

Moreover, FIG. 117(b) shows the memory cell structure using thehigh-speed plate capacitor HSPC which is adopted in one embodiment ofthe present invention. Both the STC cell and the HSPC cell are thosehaving the stereoscopic structures, in which the substrate is not usedas the electrode of the storage capacity. As a result, the memory cellshave a small area in the diffusion layer to be added to the node forstoring the information charges so that they are less deteriorated inthe information holding characteristics due to the junction leakage withthe substrate. Thanks to this memory cell, the refresh cycle number canbe increased without dropping the memory efficiency.

According to the present invention, moreover, the DRAM having a capacityas large as 16 M or more can have its power consumption reduced toreduce the heat value of the chip thereby to improve the informationholding characteristics of the memory cells.

Here, the memory cell having a small area and a stereoscopic structureis allowed to have a small diffusion layer region. As a result, theinformation holding characteristics due to the junction leakage areimproved. Here, the power consumption can be dropped by combining the√{square root over (n)} refresh cycles. The information holdingcharacteristics are improved better by the low heat value.

On the other hand, the DRAM having a capacity as large as 16 Mbits ormore consumes a high power for charging or discharging the bit lines.The resultant high heat value also deteriorates the information holdingcharacteristics. Here, the √{square root over (n)} refresh cycles areindispensable. As a result, these √{square root over (n)} refresh cyclescan drop the power consumption so that the resultant low heat valueimprove the information holding characteristics.

FIG. 118(a) shows an example in which the refresh system according tothe present invention and an in-chip voltage converter are combined.FIG. 118(b) is a diagram showing a method for charging the bit linesthrough the sense amplifier by the in-chip voltage V_(CL). The voltageconverter has to be sufficiently low in impedance for feeding the chargecurrent and accordingly has to have a large area to increase the chiparea. According to the present invention, the bit line charging currentcan be dropped to reduce the area of the voltage converter andaccordingly the chip area.

FIG. 119 shows an example of the DRAM according to the prior art usingno voltage converter.

FIGS. 120(a) and 120(b) are diagrams and a time chart showing the DRAMof a structure of Vn refresh×4 for introducing the (Ai−1) address fromthe I/O pin. In the present invention, in the case of the structure of ¼n words×4 bits, the addresses pins required are more by one than thoseof the system of the prior art (as shown in FIG. 115). This additionaladdress is taken in from the I/O pin at the same timing as that ofanother row address. The data are then inputted to or outputted from theI/O pin in a time-series manner but without any operational problem.Thus, according to the present invention, it is possible to prevent thenumber of addresses pins from being increased more than that of thesystem of the prior art.

FIG. 121(a) is a layout showing another embodiment of the presentinvention. FIG. 121(b) is a circuit diagram showing a circuit forswitching to the √{square root over (n)}/2 refresh cycles in response toonly the column address signal CAS and the before row address signalRAS. This circuit is operated in the ½·√{square root over (n)} refreshcycle when in the CAS before RAS (CBR) refresh operation but otherwisein the √{square root over (n)} refresh cycle of the present invention.Since the ratio of the refresh operation to the normal operation isabout on hundredth, the current consumption is increased like the systemof the prior art only when in the CBR refresh but can be dropped whollyon average. If, moreover, the refresh operation is accomplished in theCBR refresh, the refresh cycle number can be as small as that of theprior art.

In FIG. 121(b), letters CR designate a signal which takes the level ‘L’when in the CBR refresh to accomplish a switching between the √{squareroot over (n)} refresh and ½ Vn refresh.

FIG. 122 shows a circuit for forming sense amplifier activating signalsPL, PL, PR and PR from refresh cycle switching address signals A_(xi−1)and A_(xi−1) shown in FIG. 121(b). When the CBR decision signal CR takesthe level ‘L’, both the signals A_(xi−1) and A_(Hi−1) take the ‘H’level. As a result, two word lines are selected, and all the signals PL,PL, PR and PR are outputted so that a 2√{square root over (n)} number ofsense amplifiers are activated. In these ways, the operations of½·√{square root over (n)} refresh cycles are accomplished when in theCBR.

Although not shown, the system for switching to that of the prior artmay be selectively used when multiple bits are to be simultaneouslyaccessed to, as in the multi-bit simultaneous test mode or in theburn-in mode, or when the number of the word line selections isincreased to improve the burn-in stress duty for the word lines.

FIG. 123(a) is a diagram showing a peak current increase preventingcircuit of single-phase drive type, and FIG. 123(b) is a diagram showinga peak current increase preventing circuit of two-phase drive type. Thesense amplifier driving power switch is turned ON in response to twosignals to change the timing between the two signals in response to theswitching of the refresh system. When the CR takes ‘L’, as shown in FIG.123(a), a timing delay is caused between the P1L and P1L, and P2L andP2L. When the CR takes ‘L’, as shown in FIG. 123(b), the timing delaysbetween the P1L and P1L and between P2L and P2L grow the more. Eithersystem is switched to the refresh system of the prior art, when in theCBR refresh, to prevent the peak value of the bit line charging anddischarging current from being increased. Since, however, the bit linecharging and discharging time is elongated by this system, the cycletime in the CBR refresh is elongated. FIG. 124 shows another embodimentof the present invention.

FIG. 125 is a chip layout of the DRAM using one embodiment of thepresent invention. At the chip center portion, there are arranged thebonding pads, the voltage converters (V_(CL) or V_(DL)), the substratevoltage generator (V_(BB)) and so on. The deteriorations of the memorycell characteristics are prevented by arranging the center portion withthe V_(BB) generator, which is especially liable to inject the minoritycarriers into the substrate, namely, by keeping the V_(BB) generatorapart from the memory arrays.

FIG. 126(a) is a diagram showing the case in which the refresh cyclenumber of the present embodiment is 2√{square root over (n)}, and FIG.126(b) is a diagram showing the case in which the refresh cycle numberof the present embodiment is 4√{square root over (n)}. The row addressesare more than the column addresses.

Since a DRAM of low power consumption and large capacity can befabricated without dropping the memory efficiency according to thepresent invention, it is possible to provide the DRAM which can beeasily used by the user. If the √{square root over (n)} refresh cyclesystem is adopted in the present invention as contrary to the ½·√{squareroot over (n)} refresh cycle system of the prior art, the bit linecharging and discharging current (corresponding to 50 to 70% of thetotal current to be consumed by the chip) can be dropped to one half.

Next, with reference to FIGS. 128 and 129, the dynamic RAM having abuilt-in voltage drop circuit will be described in connection with thecontents we have investigated and devised.

The voltage drop circuit is disclosed in Japanese Patent Laid-Open No.57-061981, for example.

The dynamic RAM has one built-in voltage drop circuit, as disclosedabove, which is steadily brought into operative state by the powervoltage supplied. Therefore, the voltage drop circuit is designed tohave a relatively large current feeding capacity for warranting themaximum operation current of the dynamic RAM when this RAM is selected.Our investigations have found a problem that the standby current of thedynamic RAM in the unselected state is increased to obstruct the lowpower consumption.

The invention to be disclosed hereinafter has an object to reduce thestandby current of the dynamic RAM having the built-in voltage dropcircuit thereby to promote the low power consumption.

The aforementioned and other objects and novel features of the presentinvention will become apparent from the following description taken withreference to the accompanying drawings.

The representative of the invention to be disclosed herein will bebriefly described in the following. Specifically, a voltage drop circuitto be built in the dynamic RAM or the like is constructed to comprises:a first voltage drop circuit designed to have an ability to supply sucha relatively small current as can supply a standby current when thedynamic RAM is brought into an unselected state; and a second voltagedrop circuit designed to have an ability to supply such a relativelylarge current as can supply its operating current together with saidfirst voltage drop circuit when the dynamic RAM is brought into aselected state.

According to the above-specified means, the stand-by current of thedynamic RAM can be reduced while protecting the circuit element whichhas its breakdown voltage dropped in accordance with the highintegration, dropping the power consumption of the circuit element andwarranting the maximum operation current of the circuit element in theselected state. Thus, it is possible to promote the reduction in thepower consumption of the dynamic RAM having the built-in voltage dropcircuit.

FIG. 127 is a circuit diagram showing one embodiment of the voltage dropcircuits VD1 and VD2 to which the present invention is applied. On theother hand, FIG. 128 is a block diagram showing one embodiment of thedynamic RAM which has the voltage drop circuits VD1 and VD2 of FIG. 127built therein. With reference to these Figures, the structures of thevoltage drop circuits VD1 and VD2 and the dynamic RAM according to thisembodiment will be described together with the summary of the operationsand the features thereof. Here, the individual circuit elements of FIG.127 and the circuit elements composing each block of FIG. 128 are formedover one semiconductor substrate of single crystalline silicon or thelike, although not especially limitative, by the well-known fabricationtechnology of the semiconductor integrated circuit. In FIG. 127,moreover, the MOSFETs having their channels (or back gates) indicated byarrows are of the P-channel type such that they are discriminated fromthe N-channel MOSFETs having no arrow.

The dynamic RAM of this embodiment has a relatively large storagecapacity and has its circuit elements made remarkably fine, although notespecially limitative. As a result, the circuit elements have theirbreakdown voltages dropped so that the power voltage VCC at +5V or thelike supplied from the outside cannot be fed as it is to the internalcircuits. Therefore, the dynamic RAM of this embodiment is equipped withthe voltage drop circuit which is operative to drop the power voltageVCC to an internal power voltage Vcd of +3 V, for example, and to feedit to the internal circuits, so that the power consumption of thedynamic RAM may be dropped. In this embodiment, the aforementionedvoltage drop circuit is constructed of: a (first) voltage drop circuitVD1 designed to have an ability to supply such a relatively smallcurrent as can supply a standby current when the dynamic RAM is broughtinto an unselected state; and a (second) voltage drop circuit VD2designed to have an ability to supply such a relatively large current ascan supply its operating current together with said first voltage dropcircuit, when the dynamic RAM is brought into a selected state, andbrought selectively into an operation state in accordance with the rowaddress strobe signal RAS (i.e., start control signal). As a result, thedynamic RAM has its standby current reduced to promote its low powerconsumption.

In FIG. 128, a memory array MARY is composed of: a plurality of wordlines arranged in parallel in the vertical directions of the sameFigure; a plurality of complementary data lines arranged in parallel inthe horizontal direction; and a plurality of dynamic memory cellsarranged in a lattice form at the intersections between the word linesand the complementary data lines.

The word lines composing the memory array MARY are coupled to the rowaddress decoder RAD so that they are alternatively brought into theselected state. The row address decoder RAD is fed, although especiallylimitative, with: complementary internal address signals ax0 to axi (asthe non-inverted internal address signals ax0 and the inverted internaladdress signal ax0 will be indicated together by the complementaryinternal address signals ax0) from the row address buffer RAB; and atiming signal φx from the timing generator TG.

The row address decoder RAD is selectively brought into the operativestate when the timing signal φx is raised to the high level. In thisoperative state, the row address decoder RAD decodes the aforementionedcomplementary internal address signals ax0 to axi to bring thecorresponding word lines of the memory array MARY alternately into theselected state of the high level.

The row address buffer RAB responds to the timing signal φar fed fromthe timing generator TG, to take and hold the row address signals whichare transmitted from the address multiplexer AMX. On the basis of therow address signals, moreover, the row address buffer RAB form theaforementioned complementary internal address signals ax0 to axi andfeed them to the row address decoder RAD.

When the dynamic RAM is in the normal operation mode so that the timingsignal φref of the low level is fed from the timing generator rG,although not especially limitative, the address multiplexer AMX selectsthe X-address signals AX0 to AXi, which are fed in the time-seriesmanner through the external terminals A0 to Ai, to transmit them as theaforementioned row address signals to the row address buffer RAB. When,on the other hand, the dynamic RAM is brought into the refresh mode sothat the timing signal φref is raised to the high level, the addressmultiplexer AMX selects the refresh address signals ar0 to ari, whichare fed from the refresh address counter RFC, to transmit them as theaforementioned row address signals to the row address buffer RAB.

When the dynamic RAM is in the refresh mode, although not especiallylimitative, the refresh address counter RFC accomplishes its stepwiseadvances in accordance with the timing signal φrc fed from the timinggenerator TG. As a result, the refresh address counter RFC forms andfeeds the aforementioned refresh address signals ar0 to ari to theaddress multiplexer AMX.

On the other hand, the complementary data lines constituting the memoryarray MARY are coupled at one hand to the corresponding unit amplifierof the sense amplifier SA and at the other to the corresponding switchMOSFET of the column switch CSW.

The sense amplifier SA includes a plurality of unit amplifiers which areprovided to correspond to the individual complementary data lines of thememory array MARY. These unit amplifiers are commonly fed with thetiming signal φpa from the timing generator TG.

Each of the unit amplifiers of the sense amplifier SA is selectivelybrought into an operative state when the aforementioned timing signalφpa is raised to the high level. In this operative state, each unitamplifier amplifies the fine read signal, which is outputted through thecorresponding complementary data line from a plurality of memory cellscoupled to the selected word line of the memory array MARY, and uses itas a binary read signal at the high or low level.

The column switch CSW includes plural pairs of switch MOSFETs which areprovided to correspond to the individual complementary data lines of thememory array MARY. These switch MOSFETs are coupled at one hand to thecorresponding complementary data lines of the memory array MARY, as hasbeen described hereinbefore, and at the other alternately and commonlyto the non-inverted signal line CD and the inverted signal line CD ofthe complementary common data lines. The switch MOSFET of each pair havetheir gates commonly coupled and fed with the corresponding data lineselection signal from the column address decoder CAD.

The switch MOSFETs of each pair of the column switch CSW are selectivelyturned ON as the corresponding data line selection signals arealternately raised to the high level. As a result, the correspondingcomplementary data lines of the memory array MARY are selectivelyconnected with the aforementioned complementary common data lines CD andCD.

The column address decoder CAD is fed, although not especiallylimitative, with the complementary internal address signals ay0 to ayiof (i+1) bits from the column address buffer CAB and the timing signalφy from the timing generator TG.

The column address decoder CAD is selectively brought into an operativestate as the aforementioned timing signal φy is raised to the highlevel. In this operative state, the column address decoder CAD decodesthe aforementioned complementary internal address signals ay0 to ayi toraise the corresponding data line selection signals alternately to thehigh level. These data line selection signals are fed to thecorresponding switch MOSFETs of the column switch CSW, as has beendescribed hereinbefore.

The column address buffer CAB responds to the timing signal φac fed fromthe timing generator TG, to take and hold the Y-address signals AY0 toAYi which are fed in time-series through the external terminals A0 toAi. In accordance with these Y-address signals, moreover, the columnaddress buffer CAB forms and feeds the complementary internal addresssignals ay0 to ayi to the column address decoder CAD.

The complementary common data lines CD and CD are coupled to the datainput/output circuit I/O, although not especially limitative.

This data input/output circuit I/O includes a data input buffer and adata output buffer, although not especially limitative. Of these, thedata input buffer has its input terminal coupled to the data inputterminal Din and its output terminal coupled to the complementary commondata lines CD and CD. The data input buffer is fed with the timingsignal φw from the timing generator TG, although not especiallylimitative. On the other hand, the data output buffer has its inputterminal commonly coupled to the aforementioned complementary commondata lines CD and CD and its output terminal coupled to the data outputterminal Dout. The data output buffer is fed with the timing signal φrfrom the timing generator TG.

The data input buffer of the data input/output circuit I/O isselectively brought into an operative state when the dynamic RAM isbrought into the write mode so that the aforementioned timing signal φwis raised to the high level. In this operative state, the data inputbuffer forms the complementary write signal according to the write datafed through the data input terminal Din and feeds it to the selectedmemory cell of the memory array MARY through the complementary commondata lines CD and CD. The output of the data input buffer is broughtinto the high-impedance state, although not especially limitative, whenthe timing signal φw is dropped to the low level.

The data output buffer of the data input/output circuit I/O isselectively brought into an operative state when the dynamic RAM isbrought into the read mode so that the aforementioned timing signal Oris raised to the high level. In this operative state, the data outputbuffer further amplifies the binary read signal, which is outputtedthrough the corresponding complementary data lines and the complementarycommon data lines CD and CD from the selected memory cells of the memoryarray MARY, and feeds it out from the data output terminal Dout. Whenthe timing signal φr is dropped to the low level, although notespecially limitative, the output of the data output buffer is broughtinto the high impedance state.

The timing generator TG forms and feeds the various timing signals tothe individual circuits of the dynamic RAM on the basis of the rowaddress strobe signal RAS, the column address strobe signal CAS and thewrite enable signal WE, which are fed as the start control signals fromthe outside.

The dynamic RAM of this embodiment has the voltage drop circuits VD1 andVD2 built therein, as has been described hereinbefore. These voltagedrop circuits are fed with the power voltage VCC through the externalterminals VCC and have their output terminals coupled commonly to theinternal power voltage supply point Vcd. Here, the power voltage VCC isset at a positive level such as +5 V, although not especiallylimitative. The voltage drop circuit VD2 is further fed with the timingsignal φce from the timing generator TG.

The voltage drop circuit VD1 is basically composed, as shown in FIG.127, of a pair of N-channel MOSFETs Q11 and Q12 of difference mode,although not especially limitative. Between the drains of the MOSFETsQ11 and Q12 and the aforementioned power voltage VCC, there areconnected P-channel MOSFETs Q1 and Q2. Of these, the MOSFET Q1 has itsgate and drain coupled commonly to each other and further to the gate ofthe MOSFET Q2. As a result, the MOSFETs Q1 and Q2 are in the currentmirror mode to act as the active load for the MOSFETs Q11 and Q12.

Between the commonly coupled sources of the MOSFETs Q11 and Q12 and theground potential of the circuit, there is connected an N-channel MOSFETQ13, although not especially limitative. This MOSFET Q13 is designed tohave a relatively small conductance and has its gate,and drain coupledcommonly to each other to provide a diode mode. Thus, the aforementioneddifferential MOSFETs Q11 and Q12 are always fed with the relativelysmall operation current corresponding to the conductance of the MOSFETQ13. As a result, the voltage drop circuit VD1 is steadily held in itsoperative state on condition that it is fed with the power voltage VCC.

The commonly coupled drains of the MOSFETs Q2 and Q12 are furthercoupled to the gate of a P-channel MOSFET Q3. This MOSFET Q3 has itssource coupled to the aforementioned power voltage VCC and its draincoupled to the gate of the aforementioned MOSFET Q11 and further to theinternal power voltage supply point Vcd. Thus, the MOSFET Q3 acts as acurrent feeding MOSFET for feeding the internal power voltage Vcdsubstantially to the internal circuit and as a voltage controllingMOSFET for controlling the level of the internal power voltage Vcd whenits gate voltage is changed. In this embodiment, the MOSFET Q3 isdesigned to have a relatively small conductance. As a result, thevoltage drop circuit VD1 is made to have such a relatively low currentsupply ability as can feed the standby current in the unselected stateof the dynamic RAM, because the aforementioned MOSFET Q13 is designed tohave the relatively small conductance.

The MOSFET Q12 has its gate supplied with a predetermined referencepotential Vr. Here, the reference potential Vr is generated at aconstant voltage such as +3 V, for example, by the not-shown constantvoltage generator of the dynamic RAM, although not especiallylimitative.

From these, the differential MOSFETs Q11 and Q12 function as thedifferential amplifier using the MOSFETs Q1 and Q2 as the active loadwhen the differential power voltage VCC is supplied. At this time, thedifferential amplifier compares the internal power voltage Vcd, which issupplied to its non-inverted input terminal, i.e., the gate of theMOSFET Q11, and the reference potential, which is supplied to itsinverted input terminal, i.e., the gate of the MOSFET Q12, and enlargesthe level difference to transmit it to the non-inverted output terminal,i.e., the commonly coupled drains of the MOSFETs Q2 and Q12, i.e., thegate of the MOSFET Q3. As a result, the level of the internal powervoltage Vcd is controlled and converted to the aforementioned referencepotential Vr, i.e., +3 V.

When the level of the internal power voltage Vcd is raised to exceed thereference potential Vr, the conductance of the MOSFET Q11 is increasedwhereas the conductance of the MOSFET Q12 is decreased. As a result, theMOSFET Q3 has its gate voltage raised but its conductance decreased sothat the level of the internal power voltage Vcd is dropped. If, on theother hand, the level of the internal power voltage Vcd is dropped to avalue lower than the reference voltage Vr, the conductance of the MOSFETQ11 is decreased whereas the conductance of the MOSFET Q12 is increased.Thus, the MOSFET Q3 has its gate voltage dropped and its conductanceincreased so that the level of the internal power voltage Vcd is raised.As a result, the level of the internal power voltage Vcd is converged toand stabilized at the reference potential Vr, i.e, +3 V. Since theMOSFET Q3 is designed to have the relatively low conductance, as hasbeen described hereinbefore, the current supply ability of the voltagedrop circuit VD1 is accordingly reduced.

Next, the voltage drop circuit VD2 is basically composed, as shown inFIG. 127, of a pair of N-channel MOSFETs Q14 and Q15 of differentialmode, although not especially limitative. Between these MOSFETs Q14 andQ15 and the aforementioned power voltage VCC, respectively, there areconnected P-channel MOSFET Q4 and Q5. Of these, the MOSFET Q4 has itsgate and drain coupled commonly to each other and further to the gate ofthe MOSFET Q5. Thus, the MOSFETs Q4 and Q5 are constructed to have thecurrent mirror mode and to act as the active loads for the MOSFETs Q14and Q15.

Between the commonly coupled sources of the MOSFETs Q14 and Q15 and theground potential of the circuit, there are connected in parallelN-channel MOSFETs Q16 and Q17, although not especially limitative. Ofthese, the MOSFET Q16 has its gate fed with the aforementioned timingsignal φce, and the MOSFET Q17 has its gate and drain coupled commonlyto form a diode mode. Here, the MOSFET Q16 is designed to have arelatively large conductance, whereas the MOSFET Q16 is designed to havea relatively small conductance. On the other hand, the timing signal φceis formed, as shown partially in FIG. 127, by inverting the row addressstrobe signal RAS or the start control signal of the dynamic RAM by theinverter circuit N1 of the timing generator TG, although not especiallylimitative, so that it is selectively caused to take the high level whenthe dynamic RAM is brought into its selected state.

As a result, the differential MOSFETs Q14 and Q15 thus constructed arefed with a relatively small operation current corresponding to theconductance of the MOSFET Q17, when the dynamic RAM is in its unselectedstate, and a relatively large operation current corresponding to theconductance of the MOSFET Q16 when the dynamic RAM is in its selectedstate. Thus, the voltage drop circuit VD2 is selectively brought intoits operative state when the dynamic RAM is substantially brought intothe selected state so that the timing signal φce is given the highlevel. In the unselected state of the dynamic RAM, the relatively smalloperation current is fed to the differential MOSFETs Q14 and Q15 throughthe MOSFET Q17 so that the later-described non-inverted outputterminals, i.e, the commonly coupled drains of the MOSFETs Q5 and Q15are maintained substantially at the reference potential Vr, i.e., at +3V. As a result, the rise of the voltage drop circuit Vd2 is acceleratedat the beginning of the selected state of the dynamic RAM.

The commonly coupled drains of the MOSFETs Q5 and Q15 are furthercoupled to the gate of the P-channel MOSFET Q6. This MOSFET Q6 has itssource coupled to the aforementioned power voltage VCC and its draincoupled to the gate of the MOSFET Q14 and commonly to the aforementionedinternal power voltage supply point Vcd. As a result, the MOSFET Q6 actssubstantially as a current supply MOSFET for supplying the internalpower voltage Vcd to the internal circuits and as a voltage controlMOSFET for controlling the level of the internal power voltage Vcd whenits gate voltage is changed. In this embodiment, the MOSFET Q6 isdesigned to have a relatively large conductance. Since, moreover, theMOSFET Q16 is also designed to have a relatively large conductance, thevoltage drop circuit VD2 is made to have a relatively large currentsupply ability for supplying the operation current in the selected stateof the dynamic RAM in association with the aforementioned voltage dropcircuit VD1.

The gate of the MOSFET Q15 is fed with the aforementioned referencepotential Vr. Thus, the differential MOSFETs Q14 and Q15 functionsubstantially as a differential amplifier using the MOSFETs Q4 and Q5 asactive loads when the dynamic RAM is brought into the selected state sothat the aforementioned timing signal φce takes the high level. At thistime, the differential amplifier compares the internal power voltageVcd, which is fed to its non-inverted input terminal, i.e., to the gateof the MOSFET Q14, and the reference potential Vr, which is fed to itsinverted input terminal, i.e., the gate of the MOSFET Q15, and enlargestheir level difference to feed it to the non-inverted output terminal,i.e., the commonly coupled drains of the MOSFETs Q5 and Q15, i.e., thegate of the MOSFET Q6. As a result, the level of the internal powervoltage Vcd is controlled and converged, like the case of theaforementioned voltage drop circuit VD1, to the reference potential Vr,i.e., +3 V.

In this voltage drop circuit VD2, as has been described hereinbefore, inparallel with the aforementioned MOSFET 016, there is connected a MOSFETQ17 which has a relatively small conductance and a diode mode. Thus, theforegoing differential MOSFETs Q14 and Q15 are always fed with therelatively small operation current through the MOSFET Q17. As a matterof fact, therefore, the voltage drop circuit VD2 is steadily held in theoperative state so that the potential at its non-inverted outputterminal, i.e., at the commonly coupled drains of the MOSFETs Q5 and Q15is maintained substantially at the same level as that of the operativestate. As a result, the voltage drop circuit VD2 has its rise speeded upfaster, while reducing the standby current, than the case in which theaforementioned MOSFET Q17 is omitted to set the non-inverted outputterminal substantially at power voltage VCC. Since the MOSFETs Q16 andQ6 are designed to have the relatively large conductance, it is needlessto say that the current supply ability of the voltage drop circuit VD2in the selected state of the dynamic RAM is accordingly increased tosupply the operation current sufficiently.

As described above, the dynamic RAM of this embodiment has built thereinthe voltage drop circuit for dropping the power voltage VCC fed from theoutside, such as +5 V to +3 V, for example, to feed it as the internalpower voltage. Vcd to the internal circuits. In this embodiment, thevoltage drop circuit is composed of: the voltage drop circuit VD1designed to have the relatively small current supply ability andsteadily held in its operative state; and the voltage drop circuit VD2designed to have the relatively large current supply ability and broughtselectively into its operative state substantially in response to therow address strobe signal RAS. Thus, the dynamic RAM of the presentembodiment can have its standby current reduced and its low powerconsumption promoted, while protecting its circuit elements, which havetheir breakdown voltages dropped in accordance with the highintegration, reducing its power consumption and warranting the maximumoperation current in the selected state.

As has been disclosed in the present embodiment, the following effectscan be attained by applying the present invention to the semiconductorintegrated circuit device such as the dynamic RAM having the voltagedrop circuit built therein.

-   (1) The voltage drop circuit to be built in the dynamic RAM is    composed of the first voltage drop circuit designed to have an    ability to supply such a relatively small current as can supply a    standby current when said dynamic RAM is brought into an unselected    state; and the second voltage drop circuit designed to have an    ability to supply such a relatively large current as can supply its    operating current together with the first voltage drop circuit when    the dynamic RAM is brought into a selected state. There can be    attained an effect that the standby current of the voltage drop    circuit can be reduced.-   (2) According to the above item (1), the dynamic RAM of the present    embodiment can have its standby current reduced, while protecting    its circuit elements, which have their breakdown voltages dropped in    accordance with the high integration, reducing its power consumption    and warranting the maximum operation current in the selected state.-   (3) According to the above items (1) and (2), there can be attained    an effect that the reduction in the power consumption of the dynamic    RAM can be further promoted.

Although our present invention has bee specifically described inconnection with the embodiments thereof, it should not be limited to thespecific embodiments but can naturally be modified in various mannerswithout departing the gist thereof. For example, the MOSFET Q17 of thevoltage drop circuit VD2 may be omitted from FIG. 127. Moreover, theMOSFETs Q6 and Q16 may be replaced by a plurality of P-channel MOSFETsor N-channel MOSFETs in parallel mode. The dynamic RAM may have itsvoltage drop circuits VD1 and/or VD2 composed of a plurality of voltagedrop circuits. At this time, the plural voltage drop circuitscorresponding to the voltage drop circuit VD2 may be classified inaccordance with their applications or functions so that they may beselectively brought into operative states under optimum conditions. Theconditions for forming the timing signal φec may be exemplified invarious manners. In case the dynamic RAM has the self-refresh function,for example, the starting condition for starting the refresh circuit bya refresh timer circuit may be added to the conditions for forming theaforementioned timing signal ice. In the dynamic RAM is in aquasi-static mode, the timing signal φce may be formed in response tothe chip enable signal CE, for example. In FIG. 128, the memory arrayMARY may be composed of a plurality of memory mats or may be of theso-called “multi-bit structure”, in which memory data of plural bits aresimultaneously inputted and outputted. The values of the power voltageVCC fed from the outside and the internal power voltage Vcd are notlimited by this embodiment. Moreover, a variety of modes of embodimentscan be taken in the specific circuit structures of the voltage dropcircuits VD1 and VD2 shown in FIG. 127, the block structures of thedynamic RAM shown in FIG. 128, and the combination of the controlsignals and the address signals.

Although our invention has been described hereinbefore in case it isapplied to its background field such as the dynamic RAM, it should notbe limited to those embodiments but can be applied a variety ofsemiconductor memory devices such as a static RAM or a variety ofdigital integrated circuit devices. The present invention can also beapplied widely to a semiconductor integrated circuit device having atleast a built-in voltage drop circuit.

1. A semiconductor memory device formed on a semiconductor chip comprising: a plurality of first memory arrays; a plurality of second memory arrays; a first voltage generator; and a plurality of first bonding pads; wherein the semiconductor chip is divided into a first rectangle region, a second rectangle region, and a third rectangle region, wherein the third rectangle region is arranged between the first rectangle region and the second rectangle region, wherein the plurality of first memory arrays are formed in the first rectangle region, wherein the plurality of second memory arrays are formed in the second rectangle region, wherein the voltage generator and the plurality of first bonding pads are arranged in the third rectangle region, wherein the plurality of first bonding pads are arranged between the first rectangle region and the voltage generator; wherein no bonding pads are arranged between the first voltage generator and the plurality of second memory arras, and wherein the first voltage generator is provided at a center portion between the plurality of first memory arrays and the plurality of second memory arrays.
 2. A semiconductor memory device according to claim 1, wherein the first voltage generator is a substrate voltage generator.
 3. A semiconductor memory device according to claim 2, further comprising: a second voltage generator adapted to generate an internal voltage which is supplied to the first and second memory array; and a plurality of second bonding pads, wherein the plurality of second bonding pads are arranged between the second voltage generator and the first rectangle region, and wherein no bonding pads are arranged between the second voltage generator and the second rectangle region.
 4. A semiconductor memory device according to claim 1, further comprising: a second voltage generator adapted to generate an internal voltage which is supplied to the first and second memory array; and a plurality of second bonding pads, wherein the plurality of second bonding pads are arranged between the second voltage generator and the first rectangle region, and wherein no bonding pads are arranged between the second voltage generator and the second rectangle region.
 5. A semiconductor memory device according to claim 1, wherein the semiconductor device is DRAM.
 6. A semiconductor memory device according to claim 2, wherein the semiconductor device is DRAM.
 7. A semiconductor memory device according to claim 3, wherein the semiconductor device is DRAM.
 8. A semiconductor memory device according to claim 4, wherein the semiconductor device is DRAM.
 9. A semiconductor memory device according to claim 1, wherein the first rectangle region, the third rectangle region, and the second rectangle region are sequentially arranged in a first direction, and wherein the plurality of first bonding pads are arranged in a second direction which is across the first direction.
 10. A semiconductor memory device according to claim 9, wherein the plurality of first bonding pads are arranged in two rows in the second direction. 